Signal generation method and signal generation device

ABSTRACT

A transmission method simultaneously transmitting a first modulated signal and a second modulated signal at a common frequency performs precoding on both signals using a fixed precoding matrix and regularly changes the phase of at least one of the signals, thereby improving received data signal quality for a reception device.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is based on application No. 2010-276447 filed Dec. 10,2010 in Japan, the contents of which are hereby incorporated byreference.

TECHNICAL FIELD

The present invention relates to a signal generation method and a signalgeneration apparatus for communication using multiple antennas.

BACKGROUND ART

A MIMO (Multiple-Input, Multiple-Output) system is an example of aconventional communication system using multiple antennas. Inmulti-antenna communication, of which the MIMO system is typical,multiple transmission signals are each modulated, and each modulatedsignal is simultaneously transmitted from a different antenna in orderto increase the transmission speed of the data.

FIG. 23 illustrates a sample configuration of a transmission andreception device having two transmit antennas and two receive antennas,and using two transmit modulated signals (transmit streams). In thetransmission device, encoded data are interleaved, the interleaved dataare modulated, and frequency conversion and the like are performed togenerate transmission signals, which are then transmitted from antennas.In this case, the scheme for simultaneously transmitting differentmodulated signals from different transmit antennas at the same time andon a common frequency is a spatial multiplexing MIMO system.

In this context, Patent Literature 1 suggests using a transmissiondevice provided with a different interleaving pattern for each transmitantenna. That is, the transmission device from FIG. 23 should use twodistinct interleaving patterns performed by two interleavers (π_(a) andπ_(b)). As for the reception device, Non-Patent Literature 1 andNon-Patent Literature 2 describe improving reception quality byiteratively using soft values for the detection scheme (by the MIMOdetector of FIG. 23).

As it happens, models of actual propagation environments in wirelesscommunications include NLOS (Non Line-Of-Sight), typified by a Rayleighfading environment is representative, and LOS (Line-Of-Sight), typifiedby a Rician fading environment. When the transmission device transmits asingle modulated signal, and the reception device performs maximal ratiocombination on the signals received by a plurality of antennas and thendemodulates and decodes the resulting signals, excellent receptionquality can be achieved in a LOS environment, in particular in anenvironment where the Rician factor is large. The Rician factorrepresents the received power of direct waves relative to the receivedpower of scattered waves. However, depending on the transmission system(e.g., a spatial multiplexing MIMO system), a problem occurs in that thereception quality deteriorates as the Rician factor increases (seeNon-Patent Literature 3).

FIGS. 24A and 24B illustrate an example of simulation results of the BER(Bit Error Rate) characteristics (vertical axis: BER, horizontal axis:SNR (signal-to-noise ratio) for data encoded with LDPC (low-densityparity-check) codes and transmitted over a 2×2 (two transmit antennas,two receive antennas) spatial multiplexing MIMO system in a Rayleighfading environment and in a Rician fading environment with Ricianfactors of K=3, 10, and 16 dB. FIG. 24A gives the Max-Logapproximation-based log-likelihood ratio (Max-log APP) BERcharacteristics without iterative detection (see Non-Patent Literature 1and Non-Patent Literature 2), while FIG. 24B gives the Max-log APP BERcharacteristic with iterative detection (see Non-Patent Literature 1 andNon-Patent Literature 2) (number of iterations: five). FIGS. 24A and 24Bclearly indicate that, regardless of whether or not iterative detectionis performed, reception quality degrades in the spatial multiplexingMIMO system as the Rician factor increases. Thus, the problem ofreception quality degradation upon stabilization of the propagationenvironment in the spatial multiplexing MIMO system, which does notoccur in a conventional single-modulation signal system, is unique tothe spatial multiplexing MIMO system.

Broadcast or multicast communication is a service applied to variouspropagation environments. The radio wave propagation environment betweenthe broadcaster and the receivers belonging to the users is often a LOSenvironment. When using a spatial multiplexing MIMO system having theabove problem for broadcast or multicast communication, a situation mayoccur in which the received electric field strength is high at thereception device, but in which degradation in reception quality makesservice reception difficult. In other words, in order to use a spatialmultiplexing MIMO system in broadcast or multicast communication in boththe NLOS environment and the LOS environment, a MIMO system that offersa certain degree of reception quality is desirable.

Non-Patent Literature 8 describes a scheme for selecting a codebook usedin precoding (i.e. a precoding matrix, also referred to as a precodingweight matrix) based on feedback information from a communication party.However, Non-Patent Literature 8 does not at all disclose a scheme forprecoding in an environment in which feedback information cannot beacquired from the other party, such as in the above broadcast ormulticast communication.

On the other hand, Non-Patent Literature 4 discloses a scheme forswitching the precoding matrix over time. This scheme is applicable whenno feedback information is available. Non-Patent Literature 4 disclosesusing a unitary matrix as the precoding matrix, and switching theunitary matrix at random, but does not at all disclose a schemeapplicable to degradation of reception quality in the above-describedLOS environment. Non-Patent Literature 4 simply recites hopping betweenprecoding matrices at random. Obviously, Non-Patent Literature 4 makesno mention whatsoever of a precoding method, or a structure of aprecoding matrix, for remedying degradation of reception quality in aLOS environment.

CITATION LIST Patent Literature [Patent Literature 1]

-   International Patent Application Publication No. WO2005/050885

Non-Patent Literature [Non-Patent Literature 1]

-   “Achieving near-capacity on a multiple-antenna channel” IEEE    Transaction on communications, vol. 51, no. 3, pp. 389-399, March    2003

[Non-Patent Literature 2]

-   “Performance analysis and design optimization of LDPC-coded MIMO    OFDM systems” IEEE Trans. Signal Processing, vol. 52, no. 2, pp.    348-361, February 2004

[Non-Patent Literature 3]

-   “BER performance evaluation in 2×2 MIMO spatial multiplexing systems    under Rician fading channels” IEICE Trans. Fundamentals, vol. E91-A,    no. 10, pp. 2798-2807, October 2008

[Non-Patent Literature 4]

-   “Turbo space-time codes with time varying linear transformations”    IEEE Trans. Wireless communications, vol. 6, no. 2, pp. 486-493,    February 2007

[Non-Patent Literature 5]

-   “Likelihood function for QR-MLD suitable for soft-decision turbo    decoding and its performance” IEICE Trans. Commun., vol. E88-B, no.    1, pp. 47-57, January 2004

[Non-Patent Literature 6]

-   “A tutorial on ‘Parallel concatenated (Turbo) coding’, ‘Turbo    (iterative) decoding’ and related topics” IEICE, Technical Report    IT98-51

[Non-Patent Literature 7]

-   “Advanced signal processing for PLCs: Wavelet-OFDM” Proc. of IEEE    International symposium on ISPLC 2008, pp. 187-192, 2008

[Non-Patent Literature 8]

-   D. J. Love and R. W. Heath Jr., “Limited feedback unitary precoding    for spatial multiplexing systems” IEEE Trans. Inf. Theory, vol. 51,    no. 8, pp. 2967-2976, August 2005

[Non-Patent Literature 9]

-   DVB Document A122, Framing structure, channel coding and modulation    for a second generation digital terrestrial television broadcasting    system (DVB-T2), June 2008

[Non-Patent Literature 10]

-   L. Vangelista, N. Benvenuto, and S. Tomasin “Key technologies for    next-generation terrestrial digital television standard DVB-T2,”    IEEE Commun. Magazine, vol. 47, no. 10, pp. 146-153, October 2009

[Non-Patent Literature 11]

-   T. Ohgane, T. Nishimura, and Y. Ogawa, “Application of space    division multiplexing and those performance in a MIMO channel” IEICE    Trans. Commun., vol. E88-B, no. 5, pp. 1843-1851, May 2005

[Non-Patent Literature 12]

-   R. G. Gallager “Low-density parity-check codes,” IRE Trans. Inform.    Theory, IT-8, pp. 21-28, 1962

[Non-Patent Literature 13]

-   D. J. C. Mackay, “Good error-correcting codes based on very sparse    matrices,” IEEE Trans. Inform. Theory, vol. 45, no. 2, pp. 399-431,    March 1999.

[Non-Patent Literature 14]

-   ETSI EN 302 307, “Second generation framing structure, channel    coding and modulation systems for broadcasting, interactive    services, news gathering and other broadband satellite applications”    v.1.1.2, June 2006

[Non-Patent Literature 15]

-   Y.-L. Ueng, and C.-C. Cheng “A fast-convergence decoding method and    memory-efficient VLSI decoder architecture for irregular LDPC codes    in the IEEE 802.16e standards” IEEE VTC-2007 Fall, pp. 1255-1259

[Non-Patent Literature 16]

-   S. M. Alamouti “A simple transmit diversity technique for wireless    communications” IEEE J. Select. Areas Commun., vol. 16, no. 8, pp.    1451-1458, October 1998

[Non-Patent Literature 17]

-   V. Tarokh, H. Jafrkhani, and A. R. Calderbank “Space-time block    coding for wireless communications: Performance results” IEEE J.    Select. Areas Commun., vol. 17, no. 3, no. 3, pp. 451-460, March    1999

SUMMARY OF INVENTION Technical Problem

An object of the present invention is to provide a MIMO system thatimproves reception quality in a LOS environment.

Solution to Problem

The present invention provides a signal generation scheme forgenerating, from a plurality of baseband signals, a plurality of signalsfor transmission on a common frequency band and at a common time,comprising the steps of: performing a change of phase on each of a firstbaseband signal s1 generated from a first set of bits and a secondbaseband signal s2 generated from a second set of bits, thus generatinga first post-phase change baseband signal s1′ and a second post-phasechange baseband signal s2′; and applying weighting to the firstpost-phase change baseband signal s1′ and to the second post-phasechange baseband signal s2′ according to a predetermined matrix F, thusgenerating a first weighted signal z1 and a second weighted signal z2 asthe plurality of signals for transmission on the common frequency bandand at the common time, wherein the first weighted signal z1 and thesecond weighted signal z2 satisfy the relation: (z1, z2)^(T)=F(s1′,s2′)^(T) and the change of phase is performed on the first basebandsignal s1 and the second baseband signal s2 using a phase modificationvalue sequentially selected from among N phase modification valuecandidates, N being an integer equal to or greater than two and each ofthe N phase modification value candidates being selected at least oncewithin a predetermined period.

Also, the present invention provides a signal generation apparatus forgenerating, from a plurality of baseband signals, a plurality of signalsfor transmission on a common frequency band and at a common time,comprising: a phase changer performing a change of phase on each of afirst baseband signal s1 generated from a first set of bits and a secondbaseband signal s2 generated from a second set of bits, thus generatinga first post-phase change baseband signal s1′ and a second post-phasechange baseband signal s2′; and a weighting unit applying weighting tothe first post-phase change baseband signal s1′ and to the secondpost-phase change baseband signal s2′ according to a predeterminedmatrix F, thus generating a first weighted signal z1 and a secondweighted signal z2 as the plurality of signals for transmission on thecommon frequency band and at the common time, wherein the first weightedsignal z1 and the second weighted signal z2 satisfy the relation: (z1,z2)^(T)=F(s1′, s2′)^(T) and the change of phase is performed on thefirst baseband signal s1 and the second baseband signal s2 using a phasemodification value sequentially selected from among N phase modificationvalue candidates, N being an integer equal to or greater than two andeach of the N phase modification value candidates being selected atleast once within a predetermined period.

Advantageous Effects of Invention

According to the above structure, the present invention provides asignal generation scheme and signal generation apparatus that remedydegradation of reception quality in a LOS environment, thereby providinghigh-quality service to LOS users during broadcast or multicastcommunication.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1A illustrates an example of a transmission device in a spatialmultiplexing MIMO system, FIG. 1B illustrates an example of a receptiondevice in a signal multiplexing MIMO system.

FIG. 2 illustrates a sample frame configuration.

FIG. 3 illustrates an example of a transmission device applying a phasechanging scheme.

FIG. 4 illustrates another example of a transmission device applying aphase changing scheme.

FIG. 5A illustrates another sample frame configuration, and FIG. 5Billustrates a sample antenna configuration.

FIG. 6A illustrates a sample phase changing scheme, FIG. 6B illustratesyet another sample frame composition, and FIG. 6C illustrates anothersample phase changing scheme.

FIG. 7 illustrates a sample configuration of a reception device.

FIG. 8 illustrates a sample configuration of a signal processor in thereception device.

FIG. 9 illustrates another sample configuration of a signal processor inthe reception device.

FIG. 10 illustrates an iterative decoding scheme.

FIG. 11 illustrates sample reception conditions.

FIG. 12 illustrates a further example of a transmission device applyinga phase changing scheme.

FIG. 13 illustrates yet a further example of a transmission deviceapplying a phase changing scheme.

FIG. 14A illustrates a first sample frame configuration indicating anarrangement scheme for modulated signal z1, and FIG. 14B illustrates afirst sample frame configuration indicating an arrangement scheme formodulated signal z2.

FIG. 15A illustrates a second sample frame configuration indicating anarrangement scheme for modulated signal z1, and FIG. 15B illustrates asecond sample frame configuration indicating an arrangement scheme formodulated signal z2.

FIG. 16A illustrates a third sample frame configuration indicating anarrangement scheme for modulated signal z1, and FIG. 16B illustrates athird sample frame configuration indicating an arrangement scheme formodulated signal z2.

FIG. 17A illustrates a fourth sample frame configuration indicating anarrangement scheme for modulated signal z1, and FIG. 17B illustrates afourth sample frame configuration indicating an arrangement scheme formodulated signal z2.

FIG. 18A illustrates a fifth sample frame configuration indicating anarrangement scheme for modulated signal z1, and FIG. 18B illustrates afifth sample frame configuration indicating an arrangement scheme formodulated signal z2.

FIGS. 19A and 19B illustrate examples of a mapping scheme.

FIGS. 20A and 20B illustrate further examples of a mapping scheme.

FIG. 21 illustrates a sample configuration of a weighting unit.

FIG. 22 illustrates a sample symbol rearrangement scheme.

FIG. 23 illustrates another example of a transmission and receptiondevice in a spatial multiplexing MIMO system.

FIGS. 24A and 24B illustrate sample BER characteristics.

FIG. 25 illustrates another sample phase changing scheme.

FIG. 26 illustrates yet another sample phase changing scheme.

FIG. 27 illustrates a further sample phase changing scheme.

FIG. 28 illustrates still a further sample phase changing scheme.

FIG. 29 illustrates still yet a further sample phase changing scheme.

FIG. 30 illustrates a sample symbol arrangement for a modulated signalproviding high received signal quality.

FIG. 31 illustrates a sample frame configuration for a modulated signalproviding high received signal quality.

FIG. 32 illustrates another sample symbol arrangement for a modulatedsignal providing high received signal quality.

FIG. 33 illustrates yet another sample symbol arrangement for amodulated signal providing high received signal quality.

FIG. 34 illustrates variation in numbers of symbols and slots needed percoded block when block codes are used.

FIG. 35 illustrates variation in numbers of symbols and slots needed perpair of coded blocks when block codes are used.

FIG. 36 illustrates an overall configuration of a digital broadcastingsystem.

FIG. 37 is a block diagram illustrating a sample receiver.

FIG. 38 illustrates multiplexed data configuration.

FIG. 39 is a schematic diagram illustrating multiplexing of encoded datainto streams.

FIG. 40 is a detailed diagram illustrating a video stream as containedin a PES packet sequence.

FIG. 41 is a structural diagram of TS packets and source packets in themultiplexed data.

FIG. 42 illustrates PMT data configuration.

FIG. 43 illustrates information as configured in the multiplexed data.

FIG. 44 illustrates the configuration of stream attribute information.

FIG. 45 illustrates the configuration of a video display and audiooutput device.

FIG. 46 illustrates a sample configuration of a communications system.

FIGS. 47A and 47B illustrate a variant sample symbol arrangement for amodulated signal providing high received signal quality.

FIGS. 48A and 48B illustrate another variant sample symbol arrangementfor a modulated signal providing high received signal quality.

FIGS. 49A and 49B illustrate yet another variant sample symbolarrangement for a modulated signal providing high received signalquality.

FIGS. 50A and 50B illustrate a further variant sample symbol arrangementfor a modulated signal providing high received signal quality.

FIG. 51 illustrates a sample configuration of a transmission device.

FIG. 52 illustrates another sample configuration of a transmissiondevice.

FIG. 53 illustrates a further sample configuration of a transmissiondevice.

FIG. 54 illustrates yet a further sample configuration of a transmissiondevice.

FIG. 55 illustrates a baseband signal switcher.

DESCRIPTION OF EMBODIMENTS

Embodiments of the present invention are described below with referenceto the accompanying drawings.

Embodiment 1

The following describes, in detail, a transmission scheme, atransmission device, a reception scheme, and a reception devicepertaining to the present Embodiment.

Before beginning the description proper, an outline of transmissionschemes and decoding schemes in a conventional spatial multiplexing MIMOsystem is provided. FIG. 1A illustrates the structure of a transmissiondevice in an N_(t)×N_(r) spatial multiplexing MIMO system, and FIG. 1Billustrates the structure of a reception device in an N_(t)×N_(r)spatial multiplexing MIMO system. An information vector z is encoded andinterleaved. The encoded bit vector u=(u₁, . . . u_(Nt)) is obtained asthe interleave output. Here, u_(i)=(u_(i1), . . . u_(iM)) (where M isthe number of transmitted bits per symbol). For a transmit vector s=(s₁,. . . S_(Nt)), a received signal s_(i)=map(u_(i)) is found for transmitantenna #i. Normalizing the transmit energy, this is expressible asE{|s_(i)|²}=E_(s)/N_(t) (where E_(s) is the total energy per channel).The receive vector y=(y₁, . . . y_(Nr))^(T) is expressed in Math. 1(formula 1), below.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 1} \right\rbrack & \; \\{\begin{matrix}{y = \left( {y_{1},\ldots \mspace{14mu},y_{N_{r}}} \right)^{T}} \\{= {{H_{NtNr}s} + n}}\end{matrix}\quad} & \left( {{formula}\mspace{14mu} 1} \right)\end{matrix}$

Here, H_(NtNr) is the channel matrix, n=(n₁, . . . n_(Nr)) is the noisevector, and the average value of n_(i) is zero for independent andidentically distributed (i.i.d) complex Gaussian noise of variance σ².Based on the relationship between transmitted symbols introduced into areceiver and the received symbols, the probability distribution of thereceived vectors can be expressed as Math. 2 (formula 2), below, for amulti-dimensional Gaussian distribution.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 2} \right\rbrack & \; \\{{p\left( {yu} \right)} = {\frac{1}{\left( {2{\pi\sigma}^{2}} \right)^{N_{r}}}{\exp \left( {{- \frac{1}{2\sigma^{2}}}{{y - {{Hs}(u)}}}^{2}} \right)}}} & \left( {{formula}\mspace{14mu} 2} \right)\end{matrix}$

Here, a receiver performing iterative decoding is considered. Such areceiver is illustrated in FIG. 1 as being made up of an outersoft-in/soft-out decoder and a MIMO detector. The log-likelihood ratiovector (L-value) for FIG. 1 is given by Math. 3 (formula 3) throughMath. 5 (formula 5), as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 3} \right\rbrack & \; \\{{L(u)} = \left( {{L\left( u_{1} \right)},\ldots \mspace{14mu},{L\left( u_{N_{t}} \right)}} \right)^{T}} & \left( {{formula}\mspace{14mu} 3} \right) \\\left\lbrack {{Math}.\mspace{14mu} 4} \right\rbrack & \; \\{{L\left( u_{i} \right)} = \left( {{L\left( u_{i\; 1} \right)},\ldots \mspace{14mu},{L\left( u_{iM} \right)}} \right)} & \left( {{formula}\mspace{14mu} 4} \right) \\\left\lbrack {{Math}.\mspace{14mu} 5} \right\rbrack & \; \\{{L\left( u_{ij} \right)} = {\ln \frac{P\left( {u_{ij} = {+ 1}} \right)}{P\left( {u_{ij} = {- 1}} \right)}}} & \left( {{formula}\mspace{14mu} 5} \right)\end{matrix}$

(Iterative Detection Scheme)

The following describes the MIMO signal iterative detection performed bythe N_(t)×N_(r) spatial multiplexing MIMO system.

The log-likelihood ratio of u_(mn) is defined by Math. 6 (formula 6).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 6} \right\rbrack & \; \\{{L\left( {u_{mn}y} \right)} = {\ln \; \frac{P\left( {u_{mn} = {{+ 1}y}} \right)}{P\left( {u_{mn} = {{- 1}y}} \right)}}} & \left( {{formula}\mspace{14mu} 6} \right)\end{matrix}$

Through application of Bayes' theorem, Math. 6 (formula 6) can beexpressed as Math. 7 (formula 7).

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 7} \right\rbrack} & \; \\{\begin{matrix}{{L\left( {u_{mn}y} \right)} = {\ln \frac{{p\left( {{yu_{mn}} = {+ 1}} \right)}{{P\left( {u_{mn} = {+ 1}} \right)}/{p(y)}}}{{p\left( {{yu_{mn}} = {- 1}} \right)}{{P\left( {u_{mn} = {- 1}} \right)}/{p(y)}}}}} \\{= {{\ln \frac{P\left( {u_{mn} = {+ 1}} \right)}{P\left( {u_{mn} = {- 1}} \right)}} + {\ln \frac{p\left( {{yu_{mn}} = {+ 1}} \right)}{p\left( {{yu_{mn}} = {- 1}} \right)}}}} \\{= {{\ln \frac{P\left( {u_{mn} = {+ 1}} \right)}{P\left( {u_{mn} = {- 1}} \right)}} + {\ln \frac{\sum_{U_{{mn},{+ 1}}}{{p\left( {yu} \right)}{p\left( {uu_{mn}} \right)}}}{\sum_{U_{{mn},{- 1}}}{{p\left( {yu} \right)}{p\left( {uu_{mn}} \right)}}}}}}\end{matrix}\quad} & \left( {{formula}\mspace{14mu} 7} \right)\end{matrix}$

Note that U_(mn, ±1)={u|u_(mn)=±1}. Through the approximation lnΣa_(j)˜max ln a_(j), Math. 7 (formula 7) can be approximated as Math. 8(formula 8). The symbol ˜ is herein used to signify approximation.

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 8} \right\rbrack} & \; \\{{L\left( {u_{mn}y} \right)} \approx {{\ln \frac{P\left( {u_{mn} = {+ 1}} \right)}{P\left( {u_{mn} = {- 1}} \right)}} + {\max\limits_{{Umn},{+ 1}}\left\{ {{\ln \; {p\left( {yu} \right)}} + {P\left( {uu_{mn}} \right)}} \right\}} - {\max\limits_{{Umn},{- 1}}\left\{ {{\ln \; {p\left( {yu} \right)}} + {P\left( {uu_{mn}} \right)}} \right\}}}} & \left( {{formula}\mspace{14mu} 8} \right)\end{matrix}$

In Math. 8 (formula 8), P(u|u_(mn)) and ln P(u|u_(mn)) can be expressedas follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 9} \right\rbrack & \; \\\begin{matrix}{{P\left( {uu_{mn}} \right)} = {\prod\limits_{{({ij})} \neq {({mn})}}{P\left( u_{ij} \right)}}} \\{= {\prod\limits_{{({ij})} \neq {({mn})}}\frac{\exp \left( \frac{u_{ij}{L\left( u_{ij} \right)}}{2} \right)}{{\exp \left( \frac{L\left( u_{ij} \right)}{2} \right)} + {\exp \left( {- \frac{L\left( u_{ij} \right)}{2}} \right)}}}}\end{matrix} & \left( {{formula}\mspace{14mu} 9} \right) \\\left\lbrack {{Math}.\mspace{14mu} 10} \right\rbrack & \; \\{{\ln \; {P\left( {uu_{mn}} \right)}} = {\left( {\sum\limits_{ij}{\ln \; {P\left( u_{ij} \right)}}} \right) - {\ln \; {P\left( u_{mn} \right)}}}} & \left( {{formula}\mspace{14mu} 10} \right) \\\left\lbrack {{Math}.\mspace{14mu} 11} \right\rbrack & \; \\{\begin{matrix}{{\ln \; {P\left( u_{ij} \right)}} = {{\frac{1}{2}u_{ij}{P\left( u_{ij} \right)}} - {\ln \begin{pmatrix}{{\exp \left( \frac{L\left( u_{ij} \right)}{2} \right)} +} \\{\exp \left( {- \frac{L\left( u_{j} \right)}{2}} \right)}\end{pmatrix}}}} \\{\approx {{\frac{1}{2}u_{ij}{L\left( u_{ij} \right)}} - {\frac{1}{2}{{L\left( u_{ij} \right)}}\mspace{14mu} {for}\mspace{14mu} {{L\left( u_{ij} \right)}}}} > 2} \\{= {{\frac{L\left( u_{ij} \right)}{2}}\left( {{u_{ij}\mspace{14mu} {{sign}\left( {L\left( u_{ij} \right)} \right)}} - 1} \right)}}\end{matrix}\quad} & \left( {{formula}\mspace{14mu} 11} \right)\end{matrix}$

Note that the log-probability of the equation given in Math. 2 (formula2) can be expressed as Math. 12 (formula 12).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 12} \right\rbrack & \; \\{{\ln \; {P\left( {yu} \right)}} = {{{- \frac{N_{r}}{2}}{\ln \left( {2{\pi\sigma}^{2}} \right)}} - {\frac{1}{2\sigma^{2}}{{y - {{Hs}(u)}}}^{2}}}} & \left( {{Formula}\mspace{14mu} 12} \right)\end{matrix}$

Accordingly, given Math. 7 (formula 7) and Math. 13 (formula 13), theposterior L-value for the MAP or APP (a posteriori probability) can becan be expressed as follows.

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 13} \right\rbrack} & \; \\{{L\left( {u_{mn}y} \right)} = {\ln \frac{\sum_{U_{{mn},{+ 1}}}{\exp \left\{ {{{- \frac{1}{2\sigma^{2}}}{{y - {{Hs}(u)}}}^{2}} + {\sum\limits_{ij}{\ln \; {P\left( u_{ij} \right)}}}} \right\}}}{\sum_{U_{{mn},{- 1}}}{\exp \left\{ {{{- \frac{1}{2\sigma^{2}}}{{y - {{Hs}(u)}}}^{2}} + {\sum\limits_{ij}{\ln \; {P\left( u_{ij} \right)}}}} \right\}}}}} & \left( {{formula}\mspace{14mu} 13} \right)\end{matrix}$

This is hereinafter termed iterative APP decoding. Also, given Math. 8(formula 8) and Math. 12 (formula 12), the posterior L-value for theMax-log APP can be can be expressed as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 14} \right\rbrack & \; \\{{L\left( {u_{mn}y} \right)} \approx {{\max\limits_{{Umn},{+ 1}}\left\{ {\Psi \left( {u,y,{L(u)}} \right)} \right\}} - {\max\limits_{{Umn},{- 1}}\left\{ {\Psi \left( {u,y,{L(u)}} \right)} \right\}}}} & \left( {{formula}\mspace{14mu} 14} \right) \\\left\lbrack {{Math}.\mspace{14mu} 15} \right\rbrack & \; \\{{\Psi \left( {u,y,{L(u)}} \right)} = {{{- \frac{1}{2\sigma^{2}}}{{y - {{Hs}(u)}}}^{2}} + {\sum\limits_{ij}{\ln \; {P\left( u_{{ij}\;} \right)}}}}} & \left( {{formula}\mspace{14mu} 15} \right)\end{matrix}$

This is hereinafter referred to as iterative Max-log APP decoding. Assuch, the external information required by the iterative decoding systemis obtainable by subtracting prior input from Math. 13 (formula 13) orfrom Math. 14 (formula 14).

(System Model)

FIG. 23 illustrates the basic configuration of a system related to thefollowing explanations. The illustrated system is a 2×2 spatialmultiplexing MIMO system having an outer decoder for each of two streamsA and B. The two outer decoders perform identical LDPC encoding(Although the present example considers a configuration in which theouter encoders use LDPC codes, the outer encoders are not restricted tothe use of LDPC as the error-correcting codes. The example may also berealized using other error-correcting codes, such as turbo codes,convolutional codes, or LDPC convolutional codes. Further, while theouter encoders are presently described as individually configured foreach transmit antenna, no limitation is intended in this regard. Asingle outer encoder may be used for a plurality of transmit antennas,or the number of outer encoders may be greater than the number oftransmit antennas. The system also has interleavers (π_(a), π_(b)) foreach of the streams A and B. Here, the modulation scheme is 2^(h)-QAM(i.e., h bits transmitted per symbol).

The receiver performs iterative detection (iterative APP (or Max-logAPP) decoding) of MIMO signals, as described above. The LDPC codes aredecoded using, for example, sum-product decoding.

FIG. 2 illustrates the frame configuration and describes the symbolorder after interleaving. Here, (i_(a),j_(a)) and (i_(b),j_(b)) can beexpressed as follows.

[Math. 16]

(i _(a) ,j _(a))=π_(a)(Ω_(ia,ja) ^(a))  (formula 16)

[Math. 17]

(i _(b) ,j _(b))=π_(b)(Ω_(ib,jb) ^(a))  (formula 17)

Here, i_(a) and i_(b) represent the symbol order after interleaving,j_(a) and j_(b) represent the bit position in the modulation scheme(where j_(a), j_(b)=1, . . . h), π_(a) and π_(b) represent theinterleavers of streams A and B, and Ω^(a) _(ia,ja) and Ω^(b) _(ib,jb)represent the data order of streams A and B before interleaving. Notethat FIG. 2 illustrates a situation where i_(a)=i_(b).

(Iterative Decoding)

The following describes, in detail, the sum-product decoding used indecoding the LDPC codes and the MIMO signal iterative detectionalgorithm, both used by the receiver.

Sum-Product Decoding

A two-dimensional M×N matrix H={H_(mn)} is used as the check matrix forLDPC codes subject to decoding. For the set [1,N]={1, 2 . . . N}, thepartial sets A(m) and B(n) are defined as follows.

[Math. 18]

A(m)≡{n:H _(mn)=1}  (formula 18)

[Math. 19]

B(n)≡{m:H _(mn)=1}  (formula 19)

Here, A(m) signifies the set of column indices equal to 1 for row m ofcheck matrix H, while B(n) signifies the set of row indices equal to 1for row n of check matrix H. The sum-product decoding algorithm is asfollows.

Step A-1 (Initialization): For all pairs (m,n) satisfying H_(mn)=1, setthe prior log ratio β_(mn)=1. Set the loop variable (number ofiterations) l_(sum)=1, and set the maximum number of loops l_(sum,max).Step A-2 (Processing): For all pairs (m,n) satisfying H_(mn)=1 in theorder m=1, 2, . . . M, update the extrinsic value log ratio α_(mn) usingthe following update formula.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 20} \right\rbrack & \; \\{\alpha_{mn} = {\left( {\prod\limits_{n^{\prime} \in {{A{(m)}}{\backslash n}}}{{sign}\left( {\lambda_{n^{\prime}} + \beta_{{mn}^{\prime}}} \right)}} \right) \times {f\left( {\sum\limits_{n^{\prime} \in {{A{(m)}}{\backslash n}}}{f\left( {\lambda_{n^{\prime}} + \beta_{{mn}^{\prime}}} \right)}} \right)}}} & \left( {{formula}\mspace{14mu} 20} \right) \\\left\lbrack {{Math}.\mspace{14mu} 21} \right\rbrack & \; \\{{{sign}(x)} \equiv \left\{ \begin{matrix}1 & {x \geq 0} \\{- 1} & {x < 0}\end{matrix} \right.} & \left( {{formula}\mspace{14mu} 21} \right) \\\left\lbrack {{Math}.\mspace{14mu} 22} \right\rbrack & \; \\{{f(x)} \equiv {\ln \frac{{\exp (x)} + 1}{{\exp (x)} - 1}}} & \left( {{formula}\mspace{14mu} 22} \right)\end{matrix}$

where ƒ is the Gallager function. λ_(n) can then be computed as follows.

Step A-3 (Column Operations): For all pairs (m,n) satisfying H_(mn)=1 inthe order n=1, 2, . . . N, update the extrinsic value log ratio β_(mn)using the following update formula.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 23} \right\rbrack & \; \\{\beta_{mn} = {\sum\limits_{m^{\prime} \in {{B{(n)}}\backslash m}}\alpha_{m^{\prime}n}}} & \left( {{formula}\mspace{14mu} 23} \right)\end{matrix}$

Step A-4 (Log-likelihood Ratio Calculation): For nε[1,N], thelog-likelihood ratio L_(n) is computed as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 24} \right\rbrack & \; \\{L_{n} = {{\sum\limits_{m^{\prime} \in {{B{(n)}}\backslash m}}\alpha_{m^{\prime}n}} + \lambda_{n}}} & \left( {{formula}\mspace{14mu} 24} \right)\end{matrix}$

Step A-5 (Iteration Count): If l_(sum)<l_(sum,max), then l_(sum) isincremented and the process returns to step A-2. Sum-product decodingends when l_(sum)=l_(sum,max).

The above describes one iteration of sum-product decoding operations.Afterward, MIMO signal iterative detection is performed. The variablesm, n, α_(mn), β_(mn), λ_(n), and L_(n) used in the above explanation ofsum-product decoding operations are expressed as m_(a), n_(a), α^(a)_(mana), β^(a) _(mana), λ_(na), and L_(na) for stream A and as m_(b),n_(b), α^(b) _(mbnb), β^(b) _(mbnb), λ_(nb), and L_(nb) for stream B.

(MIMO Signal Iterative Detection)

The following describes the calculation of λ_(n) for MIMO signaliterative detection.

The following formula is derivable from Math. 1 (formula 1).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 25} \right\rbrack & \; \\{\begin{matrix}{{y(t)} = \left( {{y_{1}(t)},{y_{2}(t)}} \right)^{T}} \\{= {{{H_{22}(t)}{s(t)}} + {n(t)}}}\end{matrix}\quad} & \left( {{formula}\mspace{14mu} 25} \right)\end{matrix}$

Given the frame configuration illustrated in FIG. 2, the followingfunctions are derivable from Math. 16 (formula 16) and Math. 17 (formula17).

[Math. 26]

n _(a)=Ω_(ia,ja) ^(a)  (formula 26)

[Math. 27]

n _(b)=Ω_(ib,jb) ^(b)  (formula 27)

where n_(a),n_(b)ε[1,N]. For iteration k of MIMO signal iterativedetection, the variables λ_(na), L_(na), λ_(nb), and L_(nb) areexpressed as λ_(k,na), L_(k,na), λ_(κ,nb), and L_(k,nb).

Step B-1 (Initial Detection; k=0)

For initial wave detection, λ_(o,na) and λ_(0,nb) are calculated asfollows.

For iterative APP decoding:

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 28} \right\rbrack} & \; \\{\lambda_{0,n_{x}} = {\ln \frac{\sum_{U_{0,{n_{X} + 1}}}{\exp \left\{ {{- \frac{1}{2\sigma^{2}}}{{{y\left( i_{X} \right)} - {{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}}}^{2}} \right\}}}{\sum_{U_{0,{n_{X} - 1}}}{\exp \left\{ {{- \frac{1}{2\sigma^{2}}}{{{y\left( i_{X} \right)} - {{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}}}^{2}} \right\}}}}} & \left( {{formula}\mspace{14mu} 28} \right)\end{matrix}$

For iterative Max-log APP decoding:

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 29} \right\rbrack} & \; \\{\lambda_{0,n_{X}} = {{\max\limits_{U_{0,n_{X},{+ 1}}}\left\{ {\Psi \left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)}} \right)} \right\}} - {\max\limits_{U_{0,n_{X},{- 1}}}\left\{ {\Psi \left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)}} \right)} \right\}}}} & \left( {{formula}\mspace{14mu} 29} \right) \\{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 30} \right\rbrack} & \; \\{\mspace{79mu} {{\Psi \left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)}} \right)} = {{- \frac{1}{2\sigma^{2}}}{{{y\left( i_{X} \right)} - {{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}}}^{2}}}} & \left( {{formula}\mspace{14mu} 30} \right)\end{matrix}$

where X=a,b. Next, the iteration count for the MIMO signal iterativedetection is set to l_(mimo)=0, with the maximum iteration count beingl_(mimo,max).

Step B-2 (Iterative Detection; Iteration k): When the iteration count isk, Math. 11 (formula 11), Math. 13 (formula 13) through Math. 15(formula 15), Math. 16 (formula 16), and Math. 17 (formula 17) can beexpressed as Math. 31 (formula 31) through Math. 34 (formula 34), below.Note that (X,Y)=(a,b)(b,a).

For iterative APP decoding:

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 31} \right\rbrack} & \; \\{\lambda_{k,n_{X}} = {{L_{{k - 1},\Omega_{{iX},{jX}}^{X}}\left( u_{\Omega_{{iX},{jX}}^{X}} \right)} + {\ln \frac{\begin{matrix}{\sum_{U_{k,n_{X},{+ 1}}}\exp} \\\left\{ {{{- \frac{1}{2\sigma^{2}}}{{{y\left( i_{X} \right)} - {{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}}}^{2}} + {\rho \left( u_{\Omega_{{iX},{jX}}^{X}} \right)}} \right\}\end{matrix}}{\begin{matrix}{\sum_{U_{k,n_{X},{- 1}}}\exp} \\\left\{ {{{- \frac{1}{2\sigma^{2}}}{{{y\left( i_{X} \right)} - {{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}}}^{2}} + {\rho \left( u_{\Omega_{{iX},{jX}}^{X}} \right)}} \right\}\end{matrix}}}}} & \left( {{formula}\mspace{14mu} 31} \right) \\{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 32} \right\rbrack} & \; \\{{\rho \left( u_{\Omega_{{iX},{jX}}^{X}} \right)} = {{\sum\limits_{\underset{\gamma \neq {jX}}{\gamma = 1}}^{h}{{\frac{L_{{k - 1},\Omega_{{iX},\gamma}^{X}}\left( u_{\Omega_{{iX},\gamma}^{X}} \right)}{2}}\left( {{u_{\Omega_{{iX},\gamma}^{X}}{{sign}\left( {L_{{k - 1},\Omega_{{iX},\gamma}^{X}}\left( u_{\Omega_{{iX},\gamma}^{X}} \right)} \right)}} - 1} \right)}} + {\sum\limits_{\gamma = 1}^{h}{{\frac{L_{{k - 1},\Omega_{{iX},\gamma}^{X}}\left( u_{\Omega_{{iX},\gamma}^{X}} \right)}{2}}\left( {{u_{\Omega_{{iX},\gamma}^{Y}}{{sign}\left( {L_{{k - 1},\Omega_{{iX},\gamma}^{Y}}\left( u_{\Omega_{{iX},\gamma}^{Y}} \right)} \right)}} - 1} \right)}}}} & \left( {{formula}\mspace{14mu} 32} \right)\end{matrix}$

For iterative Max-log APP decoding:

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 33} \right\rbrack} & \; \\{\lambda_{k,n_{X}} = {{L_{{k - 1},\Omega_{{iX},{jX}}^{X}}\left( u_{\Omega_{{iX},{jX}}^{X}} \right)} + {\max\limits_{U_{k,n_{X},{+ 1}}}\left\{ {\Psi \left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)},{\rho \left( u_{\Omega_{{iX},{jX}}^{X}} \right)}} \right)} \right\}} - {\max\limits_{U_{k,n_{X},{- 1}}}\left\{ {\Psi \left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)},{\rho \left( u_{\Omega_{{iX},{jX}}^{X}} \right)}} \right)} \right\}}}} & \left( {{formula}\mspace{14mu} 33} \right) \\{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 34} \right\rbrack} & \; \\{{\Psi \left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)},{\rho \left( u_{\Omega_{{iX},{jX}}^{X}} \right)}} \right)} = {{{- \frac{1}{2\sigma^{2}}}{{{y\left( i_{X} \right)} - {{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}}}^{2}} + {\rho \left( u_{\Omega_{{iX},{jX}}^{X}} \right)}}} & \left( {{formula}\mspace{14mu} 34} \right)\end{matrix}$

Step B-3 (Iteration Count and Codeword Estimation) Ifl_(mimo)<l_(mimo,max), then l_(mimo) is incremented and the processreturns to step B-2. When l_(mimo)=l_(mimo,max), an estimated codewordis found, as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 35} \right\rbrack & \; \\{{\hat{u}}_{n_{X}} = \left\{ \begin{matrix}1 & {L_{l_{mimo},n_{X}} \geq 0} \\{- 1} & {L_{l_{mimo},n_{X}} < 0}\end{matrix} \right.} & \left( {{formula}\mspace{14mu} 35} \right)\end{matrix}$

where X=a,b.

FIG. 3 shows a sample configuration of a transmission device 300pertaining to the present Embodiment. An encoder 302A takes information(data) 301A and a frame configuration signal 313 as input (whichincludes the error-correction scheme, coding rate, block length, andother information used by the encoder 302A in error correction codes ofthe data, such that the scheme designated by the frame configurationsignal 313 is used. The error-correction scheme may be switched). Inaccordance with the frame configuration signal 313, the encoder 302Aperforms error-correction coding, such as convolutional encoding, LDPCencoding, turbo encoding or similar, and outputs encoded data 303A.

An interleaver 304A takes the encoded data 303A and the frameconfiguration signal 313 as input, performs interleaving, i.e.,rearranges the order thereof, and then outputs interleaved data 305A.(Depending on the frame configuration signal 313, the interleavingscheme may be switched.)

A mapper 306A takes the interleaved data 305A and the frameconfiguration signal 313 as input and performs modulation, such as QPSK(Quadrature Phase Shift Keying), 16-QAM (16-Quadrature AmplitudeModulation), or 64-QAM (64-quadrature Amplitude Modulation) thereon,then outputs a baseband signal 307A. (Depending on the frameconfiguration signal 313, the modulation scheme may be switched.)

FIGS. 19A and 19B illustrate an example of a QPSK modulation mappingscheme for a baseband signal made up of an in-phase component I and aquadrature component Q in the IQ plane. For example, as shown in FIG.19A, when the input data are 00, then the output is I=1.0, Q=1.0.Similarly, when the input data are 01, the output is I=−1.0, Q=1.0, andso on. FIG. 19B illustrates an example of a QPSK modulation mappingscheme in the IQ plane differing from FIG. 19A in that the signal pointsof FIG. 19A have been rotated about the origin to obtain the signalpoints of FIG. 19B. Non-Patent Literature 9 and Non-Patent Literature 10describe such a constellation rotation scheme. Alternatively, the CyclicQ Delay described in Non-Patent Literature 9 and Non-Patent Literature10 may also be adopted. An alternate example, distinct from FIGS. 19Aand 19B, is shown in FIGS. 20A and 20B, which illustrate a signal pointlayout for 16-QAM in the IQ plane. The example of FIG. 20A correspondsto FIG. 19A, while that of FIG. 20B corresponds to FIG. 19B.

An encoder 302B takes information (data) 301B and the frameconfiguration signal 313 as input (which includes the error-correctionscheme, coding rate, block length, and other information used by theencoder 302A in error correction codes of the data, such that the schemedesignated by the frame configuration signal 313 is used. Theerror-correction scheme may be switched). In accordance with the frameconfiguration signal 313, the encoder 302B performs error-correctioncoding, such as convolutional encoding, LDPC encoding, turbo encoding orsimilar, and outputs encoded data 303B.

An interleaver 304B takes the encoded data 303B and the frameconfiguration signal 313 as input, performs interleaving, i.e.,rearranges the order thereof, and outputs interleaved data 305B.(Depending on the frame configuration signal 313, the interleavingscheme may be switched.)

A mapper 306B takes the interleaved data 305B and the frameconfiguration signal 313 as input and performs modulation, such as QPSK,16-QAM, or 64-QAM thereon, then outputs a baseband signal 307B.(Depending on the frame configuration signal 313, the modulation schememay be switched.)

A signal processing scheme information generator 314 takes the frameconfiguration signal 313 as input and accordingly outputs signalprocessing scheme information 315. The signal processing schemeinformation 315 designates the fixed precoding matrix to be used, andincludes information on the pattern of phase changes used for changingthe phase.

A weighting unit 308A takes baseband signal 307A, baseband signal 307B,and the signal processing scheme information 315 as input and, inaccordance with the signal processing scheme information 315, performsweighting on the baseband signals 307A and 307B, then outputs a weightedsignal 309A. The weighting scheme is described in detail, later.

A wireless unit 310A takes weighted signal 309A as input and performsprocessing such as quadrature modulation, band limitation, frequencyconversion, amplification, and so on, then outputs transmit signal 311A.Transmit signal 311A is then output as radio waves by an antenna 312A.

A weighting unit 308B takes baseband signal 307A, baseband signal 307B,and the signal processing scheme information 315 as input and, inaccordance with the signal processing scheme information 315, performsweighting on the baseband signals 307A and 307B, then outputs weightedsignal 316B.

FIG. 21 illustrates the configuration of the weighting units 308A and308B. The area of FIG. 21 enclosed in the dashed line represents one ofthe weighting units. Baseband signal 307A is multiplied by w11 to obtainw11·s1(t), and multiplied by w21 to obtain w21·s1(t). Similarly,baseband signal 307B is multiplied by w12 to obtain w12·s2(t), andmultiplied by w22 to obtain w22·s2(t). Next, z1(t)=w11·s1(t)+w12·s2(t)and z2(t)=w21·s1(t)+w22·s22(t) are obtained. Here, as explained above,s1(t) and s2(t) are baseband signals modulated according to a modulationscheme such as BPSK (Binary Phase Shift Keying), QPSK, 8-PSK (8-PhaseShift Keying), 16-QAM, 32-QAM (32-Quadrature Amplitude Modulation),64-QAM, 256-QAM 16-APSK (16-Amplitude Phase Shift Keying) and so on.

Both weighting units perform weighting using a fixed precoding matrix.The precoding matrix uses, for example, the scheme of Math. 36 (formula36), and satisfies the conditions of Math. 37 (formula 37) or Math. 38(formula 38), all found below. However, this is only an example. Thevalue of a is not restricted to Math. 37 (formula 37) and Math. 38(formula 38), and may take on other values, e.g., α=1.

Here, the precoding matrix is:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 36} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}e^{j\; 0} & {\alpha \times e^{j\; 0}} \\{\alpha \times e^{j\; 0}} & e^{j\; \pi}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 36} \right)\end{matrix}$

In Math. 36 (formula 36), above, a may be given by:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 37} \right\rbrack & \; \\{\alpha = \frac{\sqrt{2} + 4}{\sqrt{2} + 2}} & \left( {{formula}\mspace{14mu} 37} \right)\end{matrix}$

Alternatively, in Math. 36 (formula 36), above, a may be given by:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 38} \right\rbrack & \; \\{\alpha = \frac{\sqrt{2} + 3 + \sqrt{5}}{\sqrt{2} + 3 - \sqrt{5}}} & \left( {{formula}\mspace{14mu} 38} \right)\end{matrix}$

The precoding matrix is not restricted to that of Math. 36 (formula 36),but may also be as indicated by Math. 39 (formula 39).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 39} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = \begin{pmatrix}a & b \\c & d\end{pmatrix}} & \left( {{formula}\mspace{14mu} 39} \right)\end{matrix}$

In Math. 39 (formula 39), let a=Ae^(jδ11), b=Be^(jδ12), c=Ce^(jδ21), andd=De^(jδ22). Further, one of a, b, c, and d may be zero. For example,the following configurations are possible: (1) a may be zero while b, c,and d are non-zero, (2) b may be zero while a, c, and d are non-zero,(3) c may be zero while a, b, and d are non-zero, or (4) d may be zerowhile a, b, and c are non-zero.

When any of the modulation scheme, error-correcting codes, and thecoding rate thereof are changed, the precoding matrix may also be set,changed, and fixed for use.

A phase changer 317B takes weighted signal 316B and the signalprocessing scheme information 315 as input, then regularly changes thephase of the signal 316B for output. This regular change is a change ofphase performed according to a predetermined phase changing patternhaving a predetermined period (cycle) (e.g., every n symbols (n being aninteger, n≧1) or at a predetermined interval). The details of the phasechanging pattern are explained below, in Embodiment 4.

Wireless unit 310B takes post-phase change signal 309B as input andperforms processing such as quadrature modulation, band limitation,frequency conversion, amplification, and so on, then outputs transmitsignal 311B. Transmit signal 311B is then output as radio waves by anantenna 312B.

FIG. 4 illustrates a sample configuration of a transmission device 400that differs from that of FIG. 3. The points of difference of FIG. 4from FIG. 3 are described next.

An encoder 402 takes information (data) 401 and the frame configurationsignal 313 as input, and, in accordance with the frame configurationsignal 313, performs error-correction coding and outputs encoded data402.

A distributor 404 takes the encoded data 403 as input, performsdistribution thereof, and outputs data 405A and data 405B. Although FIG.4 illustrates only one encoder, the number of encoders is not limited assuch. The present invention may also be realized using m encoders (mbeing an integer, m≧1) such that the distributor divides the encodeddata created by each encoder into two groups for distribution.

FIG. 5A illustrates an example of a frame configuration in the timedomain for a transmission device according to the present Embodiment.Symbol 500_1 is for notifying the reception device of the transmissionscheme. For example, symbol 500_1 conveys information such as theerror-correction scheme used for transmitting data symbols, the codingrate thereof, and the modulation scheme used for transmitting datasymbols.

Symbol 501_1 is for estimating channel fluctuations for modulated signalz1(t) (where t is time) transmitted by the transmission device. Symbol502_1 is a data symbol transmitted by modulated signal z1(t) as symbolnumber u (in the time domain). Symbol 503_1 is a data symbol transmittedby modulated signal z1(t) as symbol number u+1.

Symbol 501_2 is for estimating channel fluctuations for modulated signalz2(t) (where t is time) transmitted by the transmission device. Symbol502_2 is a data symbol transmitted by modulated signal z2(t) as symbolnumber u (in the time domain). Symbol 503_2 is a data symbol transmittedby modulated signal z1(t) as symbol number u+1.

Here, the symbols of z1(t) and of z2(t) having the same time (identicaltiming) are transmitted from the transmit antenna using the same(shared/common) frequency.

FIG. 5B illustrates an example of an antenna configuration of atransmission device and a reception device of the present Embodiment.

The following describes the relationships between the modulated signalsz1(t) and z2(t) transmitted by the transmission device and the receivedsignals r1(t) and r2(t) received by the reception device.

In FIGS. 5, 504#1 and 504#2 indicate transmit antennas of thetransmission device, while 505#1 and 505#2 indicate receive antennas ofthe reception device. The transmission device transmits modulated signalz1(t) from transmit antenna 504#1 and transmits modulated signal z2(t)from transmit antenna 504#2. Here, the modulated signals z1(t) and z2(t)are assumed to occupy the same (shared/common) frequency (bandwidth).The channel fluctuations in the transmit antennas of the transmissiondevice and the antennas of the reception device are h₁₁(t), h₁₂(t),h₂₁(t), and h₂₂(t), respectively. Assuming that receive antenna 505#1 ofthe reception device receives received signal r1(t) and that receiveantenna 505#2 of the reception device receives received signal r2(t),the following relationship holds.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 40} \right\rbrack & \; \\{\begin{pmatrix}{r\; 1(t)} \\{r\; 2(t)}\end{pmatrix} = {\begin{pmatrix}{h_{11}(t)} & {h_{12}(t)} \\{h_{21}(t)} & {h_{22}(t)}\end{pmatrix}\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 40} \right)\end{matrix}$

FIG. 6 pertains to the weighting scheme (precoding scheme) of thepresent Embodiment, FIG. 6B pertains to a frame configuration of thepresent Embodiment, and FIG. 6C pertains to the phase changing scheme ofthe present Embodiment. A weighting unit 600 is a combined version ofthe weighting units 308A and 308B from FIG. 3. As shown, stream s1(t)and stream s2(t) correspond to the baseband signals 307A and 307B ofFIG. 3. That is, the streams s1(t) and s2(t) are baseband signals madeup of an in-phase component I and a quadrature component Q conforming tomapping by a modulation scheme such as QPSK, 16-QAM, and 64-QAM. Asindicated by the frame configuration of FIG. 6B, stream s1(t) isrepresented as s1(u) at symbol number u, as s1(u+1) at symbol numberu+1, and so forth. Similarly, stream s2(t) is represented as s2(u) atsymbol number u, as s2(u+1) at symbol number u+1, and so forth. Theweighting unit 600 takes the baseband signals 307A (s1(t)) and 307B(s2(t)) as well as the signal processing scheme information 315 fromFIG. 3 as input, performs weighting in accordance with the signalprocessing scheme information 315, and outputs the weighted signals 309A(z1(t)) and 316B(z2′(t)) from FIG. 3. The phase changer 317B changes thephase of weighted signal 316B(z2′(t)) and outputs post-phase changesignal 309B(z2(t)).

Here, given vector W1=(w11,w12) from the first row of the fixedprecoding matrix F, z1(t) is expressible as Math. 41 (formula 41),below.

[Math. 41]

z1(t)=W1×(s1(t),s2(t))^(T)  (formula 41)

Similarly, given vector W2=(w21,w22) from the second row of the fixedprecoding matrix F, and letting the phase changing formula applied bythe phase changer by y(t), then z2(t) is expressible as Math. 42(formula 42), below.

[Math. 42]

z2(t)=y(t)×W2×(s1(t),s2(t))^(T)  (formula 42)

Here, y(t) is a phase changing formula following a predetermined scheme.For example, given a period (cycle) of four and time u, the phasechanging formula is expressible as Math. 43 (formula 43), below.

[Math. 43]

y(u)=e ^(j0)  (formula 43)

Similarly, the phase changing formula for time u+1 may be, for example,as given by Math. 44 (formula 44).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 44} \right\rbrack & \; \\{{y\left( {u + 1} \right)} = e^{j\frac{\pi}{2\;}}} & \left( {{formula}\mspace{14mu} 44} \right)\end{matrix}$

That is, the phase changing formula for time u+k is expressible as Math.45 (formula 45).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 45} \right\rbrack & \; \\{{y\left( {u + k} \right)} = e^{j\frac{k\; \pi}{2\;}}} & \left( {{formula}\mspace{14mu} 45} \right)\end{matrix}$

Note that Math. 43 (formula 43) through Math. 45 (formula 45) are givenonly as an example of regular phase changing.

The regular change of phase is not restricted to a period (cycle) offour. Improved reception capabilities (the error-correctioncapabilities, to be exact) may potentially be promoted in the receptiondevice by increasing the period (cycle) number (this does not mean thata greater period (cycle) is better, though avoiding small numbers suchas two is likely ideal).

Furthermore, although Math. 43 (formula 43) through Math. 45 (formula45), above, represent a configuration in which a change in phase iscarried out through rotation by consecutive predetermined phases (in theabove formula, every π/2), the change in phase need not be rotation by aconstant amount, but may also be random. For example, in accordance withthe predetermined period (cycle) of y(t), the phase may be changedthrough sequential multiplication as shown in Math. 46 (formula 46) andMath. 47 (formula 47). The key point of regular phase changing is thatthe phase of the modulated signal is regularly changed. The degree ofphase change is preferably as even as possible, such as from −π radiansto π radians. However, given that this describes a distribution, randomchanges are also possible.

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 46} \right\rbrack} & \; \\\left. e^{j\; 0}\rightarrow\left. e^{j\; \frac{\pi}{5}}\rightarrow\left. e^{j\; \frac{2\pi}{5}}\rightarrow\left. e^{j\; \frac{3\pi}{5}}\rightarrow\left. e^{j\; \frac{4\pi}{5}}\rightarrow\left. e^{j\; \frac{2\pi}{5}}\rightarrow\left. e^{j\; \pi}\rightarrow\left. e^{j\; \frac{6\pi}{5}}\rightarrow\left. e^{j\; \frac{7\pi}{5}}\rightarrow\left. e^{j\; \frac{8\pi}{5}}\rightarrow e^{j\; \frac{9\pi}{5}} \right. \right. \right. \right. \right. \right. \right. \right. \right. \right. & \left( {{formula}\mspace{14mu} 46} \right) \\{\mspace{79mu} \left\lbrack {{Math}.\mspace{14mu} 47} \right\rbrack} & \; \\\left. e^{j\; \frac{\pi}{2}}\rightarrow\left. e^{j\; \pi}\rightarrow\left. e^{j\; \frac{3\pi}{2}}\rightarrow\left. e^{j\; 2\pi}\rightarrow\left. e^{j\; \frac{\pi}{4}}\rightarrow\left. e^{j\; \frac{3}{4}\pi}\rightarrow\left. e^{j\; \frac{5\pi}{4}}\rightarrow e^{j\; \frac{7\pi}{4}} \right. \right. \right. \right. \right. \right. \right. & \left( {{formula}\mspace{14mu} 47} \right)\end{matrix}$

As such, the weighting unit 600 of FIG. 6 performs precoding usingfixed, predetermined precoding weights, and the phase changer 317Bchanges the phase of the signal input thereto while regularly varyingthe phase changing degree.

When a specialized precoding matrix is used in a LOS environment, thereception quality is likely to improve tremendously. However, dependingon the direct wave conditions, the phase and amplitude components of thedirect wave may greatly differ from the specialized precoding matrix,upon reception. The LOS environment has certain rules. Thus, datareception quality is tremendously improved through a regular changeapplied to a transmit signal that obeys those rules. The presentinvention offers a signal processing scheme for improvements in the LOSenvironment.

FIG. 7 illustrates a sample configuration of a reception device 700pertaining to the present embodiment. Wireless unit 703_X receives, asinput, received signal 702_X received by antenna 701_X, performsprocessing such as frequency conversion, quadrature demodulation, andthe like, and outputs baseband signal 704_X.

Channel fluctuation estimator 705_1 for modulated signal z1 transmittedby the transmission device takes baseband signal 704_X as input,extracts reference symbol 501_1 for channel estimation from FIG. 5,estimates the value of h₁₁ from Math. 40 (formula 40), and outputschannel estimation signal 706_1.

Channel fluctuation estimator 705_2 for modulated signal z2 transmittedby the transmission device takes baseband signal 704_X as input,extracts reference symbol 501_2 for channel estimation from FIG. 5,estimates the value of h₁₂ from Math. 40 (formula 40), and outputschannel estimation signal 706_2.

Wireless unit 703_Y receives, as input, received signal 702_Y receivedby antenna 701_X, performs processing such as frequency conversion,quadrature demodulation, and the like, and outputs baseband signal704_Y.

Channel fluctuation estimator 707_1 for modulated signal z1 transmittedby the transmission device takes baseband signal 704_Y as input,extracts reference symbol 501_1 for channel estimation from FIG. 5,estimates the value of h₂₁ from Math. 40 (formula 40), and outputschannel estimation signal 708_1.

Channel fluctuation estimator 707_2 for modulated signal z2 transmittedby the transmission device takes baseband signal 704_Y as input,extracts reference symbol 501_2 for channel estimation from FIG. 5,estimates the value of h₂₂ from Math. 40 (formula 40), and outputschannel estimation signal 708_2.

A control information decoder 709 receives baseband signal 704_X andbaseband signal 704_Y as input, detects symbol 500_1 that indicates thetransmission scheme from FIG. 5, and outputs a transmission schemeinformation signal 710 for the transmission device.

A signal processor 711 takes the baseband signals 704_X and 704_Y, thechannel estimation signals 706_1, 706_2, 708_1, and 708_2, and thetransmission scheme information signal 710 as input, performs detectionand decoding, and then outputs received data 712_1 and 712_2.

Next, the operations of the signal processor 711 from FIG. 7 aredescribed in detail. FIG. 8 illustrates a sample configuration of thesignal processor 711 pertaining to the present embodiment. As shown, thesignal processor 711 is primarily made up of an inner MIMO detector,soft-in/soft-out decoders, and a coefficient generator. Non-PatentLiterature 2 and Non-Patent Literature 3 describe a scheme of iterativedecoding using this structure. The MIMO system described in Non-PatentLiterature 2 and Non-Patent Literature 3 is a spatial multiplexing MIMOsystem, while the present Embodiment differs from Non-Patent Literature2 and Non-Patent Literature 3 in describing a MIMO system that regularlychanges the phase over time while using the same precoding matrix.Taking the (channel) matrix H(t) of Math. 36 (formula 36), then byletting the precoding weight matrix from FIG. 6 be F (here, a fixedprecoding matrix remaining unchanged for a given received signal) andletting the phase changing formula used by the phase changer from FIG. 6be Y(t) (here, Y(t) changes over time t), then the receive vectorR(t)=(r1(t),r2(t))^(T) and the stream vector S(t)=(s1(t),s2(t))^(T) thefollowing function is derived:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 48} \right\rbrack & \; \\{{{R(t)} = {{H(t)} \times {Y(t)} \times F \times {S(t)}}}{where}{{Y(t)} = \begin{pmatrix}1 & 0 \\0 & (t)\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 48} \right)\end{matrix}$

Here, the reception device may use the decoding schemes of Non-PatentLiterature 2 and 3 on R(t) by computing H(t)×Y(t)×F.

Accordingly, the coefficient generator 819 from FIG. 8 takes atransmission scheme information signal 818 (corresponding to 710 fromFIG. 7) indicated by the transmission device (information for specifyingthe fixed precoding matrix in use and the phase changing pattern usedwhen the phase is changed) and outputs a signal processing schemeinformation signal 820.

The inner MIMO detector 803 takes the signal processing schemeinformation signal as input and performs iterative detection anddecoding using the signal and the relationship thereof to Math. 48(formula 48). The operations thereof are described below.

The processing unit illustrated in FIG. 8 uses a processing scheme, asillustrated by FIG. 10, to perform iterative decoding (iterativedetection). First, detection of one codeword (or one frame) of modulatedsignal (stream) s1 and of one codeword (or one frame) of modulatedsignal (stream) s2 is performed. As a result, the soft-in/soft-outdecoder obtains the log-likelihood ratio of each bit of the codeword (orframe) of modulated signal (stream) s1 and of the codeword (or frame) ofmodulated signal (stream) s2. Next, the log-likelihood ratio is used toperform a second round of detection and decoding. These operations areperformed multiple times (these operations are hereinafter referred toas iterative decoding (iterative detection)). The following explanationscenter on the creation scheme of the log-likelihood ratio of a symbol ata specific time within one frame.

In FIG. 8, a memory 815 takes baseband signal 801X (corresponding tobaseband signal 704_X from FIG. 7), channel estimation signal group 802X(corresponding to channel estimation signals 706_1 and 706_2 from FIG.7), baseband signal 801Y (corresponding to baseband signal 704_Y fromFIG. 7), and channel estimation signal group 802Y (corresponding tochannel estimation signals 708_1 and 708_2 from FIG. 7) as input,executes (computes) H(t)×Y(t)×F from Math. 48 (formula 48) in order toperform iterative decoding (iterative detection) and stores theresulting matrix as a transformed channel signal group. The memory 815then outputs the above-described signals as needed, specifically asbaseband signal 816X, transformed channel estimation signal group 817X,baseband signal 816Y, and transformed channel estimation signal group817Y.

Subsequent operations are described separately for initial detection andfor iterative decoding (iterative detection).

(Initial Detection)

The inner MIMO detector 803 takes baseband signal 801X, channelestimation signal group 802X, baseband signal 801Y, and channelestimation signal group 802Y as input. Here, the modulation scheme formodulated signal (stream) s1 and modulated signal (stream) s2 is takento be 16-QAM.

The inner MIMO detector 803 first computes H(t)×Y(t)×F from the channelestimation signal groups 802X and 802Y, thus calculating a candidatesignal point corresponding to baseband signal 801X. FIG. 11 representssuch a calculation. In FIG. 11, each black dot is a candidate signalpoint in the IQ plane. Given that the modulation scheme is 16-QAM, 256candidate signal points exist. (However, FIG. 11 is only arepresentation and does not indicate all 256 candidate signal points.)Letting the four bits transmitted in modulated signal s1 be b0, b1, b2,and b3 and the four bits transmitted in modulated signal s2 be b4, b5,b6, and b7, candidate signal points corresponding to (b0, b1, b2, b3,b4, b5, b6, b7) are found in FIG. 11. The Euclidean squared distancebetween each candidate signal point and each received signal point 1101(corresponding to baseband signal 801X) is then computed. The Euclidiansquared distance between each point is divided by the noise variance σ².Accordingly, E_(X)(b0, b1, b2, b3, b4, b5, b6, b7) is calculated. Thatis, E_(X) is the Euclidian squared distance between a candidate signalpoint corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) and a receivedsignal point, divided by the noise variance. Here, each of the basebandsignals and the modulated signals s1 and s2 is a complex signal.

Similarly, the inner MIMO detector 803 computes H(t)×Y(t)×F from thechannel estimation signal groups 802X and 802Y, calculates candidatesignal points corresponding to baseband signal 801Y, computes theEuclidean squared distance between each of the candidate signal pointsand the received signal points (corresponding to baseband signal 801Y),and divides the Euclidean squared distance by the noise variance σ².Accordingly, E_(Y)(b0, b1, b2, b3, b4, b5, b6, b7) is calculated. Thatis, E_(Y) is the Euclidian squared distance between a candidate signalpoint corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) and a receivedsignal point, divided by the noise variance.

Next, E_(X)(b0, b1, b2, b3, b4, b5, b6, b7)+E_(Y)(b0, b1, b2, b3, b4,b5, b6, b7)=E(b0, b1, b2, b3, b4, b5, b6, b7) is computed.

The inner MIMO detector 803 outputs E(b0, b1, b2, b3, b4, b5, b6, b7) asa signal 804.

Log-likelihood calculator 805A takes the signal 804 as input, calculatesthe log-likelihood of bits b0, b1, b2, and b3, and outputslog-likelihood signal 806A. Note that this log-likelihood calculationproduces the log-likelihood of a bit being 1 and the log-likelihood of abit being 0. The calculation scheme is as shown in Math. 28 (formula28), Math. 29 (formula 29), and Math. 30 (formula 30), and the detailsare given by Non-Patent Literature 2 and 3.

Similarly, log-likelihood calculator 805A takes the signal 804 as input,calculates the log-likelihood of bits b0, b1, b2, and b3, and outputslog-likelihood signal 806B. A deinterleaver (807A) takes log-likelihoodsignal 806A as input, performs deinterleaving corresponding to that ofthe interleaver (the interleaver (304A) from FIG. 3), and outputsdeinterleaved log-likelihood signal 808A.

Similarly, a deinterleaver (807B) takes log-likelihood signal 806B asinput, performs deinterleaving corresponding to that of the interleaver(the interleaver (304B) from FIG. 3), and outputs deinterleavedlog-likelihood signal 808B.

Log-likelihood ratio calculator 809A takes deinterleaved log-likelihoodsignal 808A as input, calculates the log-likelihood ratio of the bitsencoded by encoder 302A from FIG. 3, and outputs log-likelihood ratiosignal 810A.

Similarly, log-likelihood ratio calculator 809B takes deinterleavedlog-likelihood signal 808B as input, calculates the log-likelihood ratioof the bits encoded by encoder 302B from FIG. 3, and outputslog-likelihood ratio signal 810B.

Soft-in/soft-out decoder 811A takes log-likelihood ratio signal 810A asinput, performs decoding, and outputs decoded log-likelihood ratio 812A.

Similarly, soft-in/soft-out decoder 811B takes log-likelihood ratiosignal 810B as input, performs decoding, and outputs decodedlog-likelihood ratio 812B.

(Iterative Decoding (Iterative Detection), k Iterations)

The interleaver (813A) takes the k−1th decoded log-likelihood ratio 812Adecoded by the soft-in/soft-out decoder as input, performs interleaving,and outputs interleaved log-likelihood ratio 814A. Here, theinterleaving pattern used by the interleaver (813A) is identical to thatof the interleaver (304A) from FIG. 3.

Another interleaver (813B) takes the k−1th decoded log-likelihood ratio812B decoded by the soft-in/soft-out decoder as input, performsinterleaving, and outputs interleaved log-likelihood ratio 814B. Here,the interleaving pattern used by the other interleaver (813B) isidentical to that of another interleaver (304B) from FIG. 3.

The inner MIMO detector 803 takes baseband signal 816X, transformedchannel estimation signal group 817X, baseband signal 816Y, transformedchannel estimation signal group 817Y, interleaved log-likelihood ratio814A, and interleaved log-likelihood ratio 814B as input. Here, basebandsignal 816X, transformed channel estimation signal group 817X, basebandsignal 816Y, and transformed channel estimation signal group 817Y areused instead of baseband signal 801X, channel estimation signal group802X, baseband signal 801Y, and channel estimation signal group 802Ybecause the latter cause delays due to the iterative decoding.

The iterative decoding operations of the inner MIMO detector 803 differfrom the initial detection operations thereof in that the interleavedlog-likelihood ratios 814A and 814B are used in signal processing forthe former. The inner MIMO detector 803 first calculates E(b0, b1, b2,b3, b4, b5, b6, b7) in the same manner as for initial detection. Inaddition, the coefficients corresponding to Math. 11 (formula 11) andMath. 32 (formula 32) are computed from the interleaved log-likelihoodratios 814A and 814B. The value of E(b0, b1, b2, b3, b4, b5, b6, b7) iscorrected using the coefficients so calculated to obtain E′(b0, b1, b2,b3, b4, b5, b6, b7), which is output as the signal 804.

Log-likelihood calculator 805A takes the signal 804 as input, calculatesthe log-likelihood of bits b0, b1, b2, and b3, and outputs thelog-likelihood signal 806A. Note that this log-likelihood calculationproduces the log-likelihood of a bit being 1 and the log-likelihood of abit being 0. The calculation scheme is as shown in Math. 31 (formula 31)through Math. 35 (formula 35), and the details are given by Non-PatentLiterature 2 and 3.

Similarly, log-likelihood calculator 805B takes the signal 804 as input,calculates the log-likelihood of bits b4, b5, b6, and b7, and outputsthe log-likelihood signal 806A. Operations performed by thedeinterleaver onwards are similar to those performed for initialdetection.

While FIG. 8 illustrates the configuration of the signal processor whenperforming iterative detection, this structure is not absolutelynecessary as good reception improvements are obtainable by iterativedetection alone. As long as the components needed for iterativedetection are present, the configuration need not include theinterleavers 813A and 813B. In such a case, the inner MIMO detector 803does not perform iterative detection.

The key point for the present Embodiment is the calculation ofH(t)×Y(t)×F. As shown in Non-Patent Literature 5 and the like, QRdecomposition may also be used to perform initial detection anditerative detection.

Also, as indicated by Non-Patent Literature 11, MMSE (MinimumMean-Square Error) and ZF (Zero-Forcing) linear operations may beperformed based on H(t)×Y(t)×F when performing initial detection.

FIG. 9 illustrates the configuration of a signal processor, unlike thatof FIG. 8, that serves as the signal processor for modulated signalstransmitted by the transmission device from FIG. 4. The point ofdifference from FIG. 8 is the number of soft-in/soft-out decoders. Asoft-in/soft-out decoder 901 takes the log-likelihood ratio signals 810Aand 810B as input, performs decoding, and outputs a decodedlog-likelihood ratio 902. A distributor 903 takes the decodedlog-likelihood ratio 902 as input for distribution. Otherwise, theoperations are identical to those explained for FIG. 8.

As described above, when a transmission device according to the presentEmbodiment using a MIMO system transmits a plurality of modulatedsignals from a plurality of antennas, changing the phase over time whilemultiplying by the precoding matrix so as to regularly change the phaseresults in improvements to data reception quality for a reception devicein a LOS environment where direct waves are dominant, in contrast to aconventional spatial multiplexing MIMO system.

In the present Embodiment, and particularly in the configuration of thereception device, the number of antennas is limited and explanations aregiven accordingly. However, the Embodiment may also be applied to agreater number of antennas. In other words, the number of antennas inthe reception device does not affect the operations or advantageouseffects of the present Embodiment.

Also, although LDPC codes are described as a particular example, thepresent Embodiment is not limited in this manner. Furthermore, thedecoding scheme is not limited to the sum-product decoding example givenfor the soft-in/soft-out decoder. Other soft-in/soft-out decodingschemes, such as the BCJR algorithm, SOVA, and the Max-Log-Map algorithmmay also be used. Details are provided in Non-Patent Literature 6.

In addition, although the present Embodiment is described using asingle-carrier scheme, no limitation is intended in this regard. Thepresent Embodiment is also applicable to multi-carrier transmission.Accordingly, the present Embodiment may also be realized using, forexample, spread-spectrum communications, OFDM (OrthogonalFrequency-Division Multiplexing), SC-FDMA (Single CarrierFrequency-Division Multiple Access), SC-OFDM (Single Carrier OrthogonalFrequency-Division Multiplexing), wavelet OFDM as described inNon-Patent Literature 7, and so on. Furthermore, in the presentEmbodiment, symbols other than data symbols, such as pilot symbols(preamble, unique word, etc) or symbols transmitting controlinformation, may be arranged within the frame in any manner.

The following describes an example in which OFDM is used as amulti-carrier scheme.

FIG. 12 illustrates the configuration of a transmission device usingOFDM. In FIG. 12, components operating in the manner described for FIG.3 use identical reference numbers.

OFDM-related processor 1201A takes weighted signal 309A as input,performs OFDM-related processing thereon, and outputs transmit signal1202A. Similarly, OFDM-related processor 1201B takes post-phase change309B as input, performs OFDM-related processing thereon, and outputstransmit signal 1202A

FIG. 13 illustrates a sample configuration of the OFDM-relatedprocessors 1201A and 1201B and onward from FIG. 12. Components 1301Athrough 1310A belong between 1201A and 312A from FIG. 12, whilecomponents 1301B through 1310B belong between 1201B and 312B.

Serial-to-parallel converter 1302A performs serial-to-parallelconversion on weighted signal 1301A (corresponding to weighted signal309A from FIG. 12) and outputs parallel signal 1303A.

Reorderer 1304A takes parallel signal 1303A as input, performsreordering thereof, and outputs reordered signal 1305A. Reordering isdescribed in detail later.

IFFT (Inverse Fast Fourier Transform) unit 1306A takes reordered signal1305A as input, applies an IFFT thereto, and outputs post-IFFT signal1307A.

Wireless unit 1308A takes post-IFFT signal 1307A as input, performsprocessing such as frequency conversion and amplification, thereon, andoutputs modulated signal 1309A. Modulated signal 1309A is then output asradio waves by antenna 1310A.

Serial-to-parallel converter 1302B performs serial-to-parallelconversion on weighted signal 1301B (corresponding to post-phase change309B from FIG. 12) and outputs parallel signal 1303B.

Reorderer 1304B takes parallel signal 1303B as input, performsreordering thereof, and outputs reordered signal 1305B. Reordering isdescribed in detail later.

IFFT unit 1306B takes reordered signal 1305B as input, applies an IFFTthereto, and outputs post-IFFT signal 1307B.

Wireless unit 1308B takes post-IFFT signal 1307B as input, performsprocessing such as frequency conversion and amplification thereon, andoutputs modulated signal 1309B. Modulated signal 1309B is then output asradio waves by antenna 1310A.

The transmission device from FIG. 3 does not use a multi-carriertransmission scheme. Thus, as shown in FIG. 6, the change of phase isperformed to achieve a period (cycle) of four and the post-phase changesymbols are arranged with respect to the time domain. As shown in FIG.12, when multi-carrier transmission, such as OFDM, is used, then,naturally, precoded post-phase change symbols may be arranged withrespect to the time domain as in FIG. 3, and this applies to each(sub-)carrier. However, for multi-carrier transmission, the arrangementmay also be in the frequency domain, or in both the frequency domain andthe time domain. The following describes these arrangements.

FIGS. 14A and 14B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13.The frequency axes are made up of (sub-)carriers 0 through 9. Themodulated signals z1 and z2 share common times (timing) and use a commonfrequency band. FIG. 14A illustrates a first example of a frameconfiguration indicating a reordering scheme for the symbols ofmodulated signal z1, while FIG. 14B illustrates a first example of aframe configuration indicating a reordering scheme for the symbols ofmodulated signal z2. With respect to the symbols of weighted signal1301A input to serial-to-parallel converter 1302A, the assigned orderingis #0, #1, #2, #3, and so on. Here, given that the example deals with aperiod (cycle) of four, #0, #1, #2, and #3 are equivalent to one period(cycle). Similarly, #4n, #4n+1, #4n+2, and #4n+3 (n being a non-zeropositive integer) are also equivalent to one period (cycle).

As shown in FIG. 14A, symbols #0, #1, #2, #3, and so on are arranged inorder, beginning at carrier 0. Symbols #0 through #9 are given time $1,followed by symbols #10 through #19 which are given time #2, and so onin a regular arrangement. Note that the modulated signals z1 and z2 arecomplex signals.

Similarly, with respect to the symbols of weighted signal 1301B input toserial-to-parallel converter 1302B, the assigned ordering is #0, #1, #2,#3, and so on. Here, given that the example deals with a period (cycle)of four, a different change of phase is applied to each of #0, #1, #2,and #3, which are equivalent to one period (cycle). Similarly, adifferent change of phase is applied to each of #4n, #4n+1, #4n+2, and#4n+3 (n being a non-zero positive integer), which are also equivalentto one period (cycle)

As shown in FIG. 14B, symbols #0, #1, #2, #3, and so on are arranged inorder, beginning at carrier 0. Symbols #0 through #9 are given time $1,followed by symbols #10 through #19 which are given time #2, and so onin a regular arrangement.

The symbol group 1402 shown in FIG. 14B corresponds to one period(cycle) of symbols when the phase changing scheme of FIG. 6A, FIG. 6B,and FIG. 6C is used. Symbol #0 is the symbol obtained by using the phaseat time u in FIG. 6C, symbol #1 is the symbol obtained by using thephase at time u+1 in FIG. 6C, symbol #2 is the symbol obtained by usingthe phase at time u+2 in FIG. 6C, and symbol #3 is the symbol obtainedby using the phase at time u+3 in FIG. 6C. Accordingly, for any symbol#x, symbol #x is the symbol obtained by using the phase at time u inFIG. 6C when x mod 4 equals 0 (i.e., when the remainder of x divided by4 is 0, mod being the modulo operator), symbol #x is the symbol obtainedby using the phase at time u+1 in FIG. 6 when x mod 4 equals 1, symbol#x is the symbol obtained by using the phase at time u+2 in FIG. 6 whenx mod 4 equals 2, and symbol #x is the symbol obtained by using thephase at time u+3 in FIG. 6C when x mod 4 equals 3.

In the present Embodiment, modulated signal z1 shown in FIG. 14A has notundergone a change of phase.

As such, when using a multi-carrier transmission scheme such as OFDM,and unlike single carrier transmission, symbols may be arranged withrespect to the frequency domain. Of course, the symbol arrangementscheme is not limited to those illustrated by FIGS. 14A and 14B. Furtherexamples are shown in FIGS. 15A, 15B, 16A, and 16B.

FIGS. 15A and 15B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 14A and 14B. FIG. 15A illustrates secondexample of a frame configuration indicating a reordering scheme for thesymbols of modulated signal z1, while FIG. 15B illustrates a secondexample of a frame configuration indicating a reordering scheme for thesymbols of modulated signal z2. FIGS. 15A and 15B differ from FIGS. 14Aand 14B in that different reordering schemes are applied to the symbolsof modulated signal z1 and to the symbols of modulated signal z2. InFIG. 15B, symbols #0 through #5 are arranged at carriers 4 through 9,symbols #6 though #9 are arranged at carriers 0 through 3, and thisarrangement is repeated for symbols #10 through #19. Here, as in FIG.14B, symbol group 1502 shown in FIG. 15B corresponds to one period(cycle) of symbols when the phase changing scheme of FIGS. 6A through 6Cis used.

FIGS. 16A and 16B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 14A and 14B. FIG. 16A illustrates athird example of a frame configuration indicating a reordering schemefor the symbols of modulated signal z1, while FIG. 16B illustrates athird example of a frame configuration indicating a reordering schemefor the symbols of modulated signal z2. FIGS. 16A and 16B differ fromFIGS. 14A and 14B in that, while FIGS. 14A and 14B showed symbolsarranged at sequential carriers, FIGS. 16A and 16B do not arrange thesymbols at sequential carriers. Obviously, for FIGS. 16A and 16B,different reordering schemes may be applied to the symbols of modulatedsignal z1 and to the symbols of modulated signal z2 as in FIGS. 15A and15B.

FIGS. 17A and 17B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from those of FIGS. 14A through 16B. FIG. 17A illustrates afourth example of a frame configuration indicating a reordering schemefor the symbols of modulated signal z1 and FIG. 17B illustrates a fourthexample of a frame configuration indicating a reordering scheme for thesymbols of modulated signal z2. While FIGS. 14A through 16B show symbolsarranged with respect to the frequency axis, FIGS. 17A and 17B use thefrequency and time axes together in a single arrangement.

While FIG. 6 describes an example where a change of phase is performedin a four slot period (cycle), the following example describes an eightslot period (cycle). In FIGS. 17A and 17B, the symbol group 1702 isequivalent to one period (cycle) of symbols when the phase changingscheme is used (i.e., to eight symbols) such that symbol #0 is thesymbol obtained by using the phase at time u, symbol #1 is the symbolobtained by using the phase at time u+1, symbol #2 is the symbolobtained by using the phase at time u+2, symbol #3 is the symbolobtained by using the phase at time u+3, symbol #4 is the symbolobtained by using the phase at time u+4, symbol #5 is the symbolobtained by using the phase at time u+5, symbol #6 is the symbolobtained by using the phase at time u+6, and symbol #7 is the symbolobtained by using the phase at time u+7. Accordingly, for any symbol #x,symbol #x is the symbol obtained by using the phase at time u when x mod8 equals 0, symbol #x is the symbol obtained by using the phase at timeu+1 when x mod 8 equals 1, symbol #x is the symbol obtained by using thephase at time u+2 when x mod 8 equals 2, symbol #x is the symbolobtained by using the phase at time u+3 when x mod 8 equals 3, symbol #xis the symbol obtained by using the phase at time u+4 when x mod 8equals 4, symbol #x is the symbol obtained by using the phase at timeu+5 when x mod 8 equals 5, symbol #x is the symbol obtained by using thephase at time u+6 when x mod 8 equals 6, and symbol #x is the symbolobtained by using the phase at time u+7 when x mod 8 equals 7. In FIGS.17A and 17B four slots along the time axis and two slots along thefrequency axis are used for a total of 4×2=8 slots, in which one period(cycle) of symbols is arranged. Here, given m×n symbols per period(cycle) (i.e., m×n different phases are available for multiplication),then n slots (carriers) in the frequency domain and m slots in the timedomain should be used to arrange the symbols of each period (cycle),such that m>n. This is because the phase of direct waves fluctuatesslowly in the time domain relative to the frequency domain. Accordingly,the present Embodiment performs a regular change of phase that reducesthe influence of steady direct waves. Thus, the phase changing period(cycle) should preferably reduce direct wave fluctuations. Accordingly,m should be greater than n. Taking the above into consideration, usingthe time and frequency domains together for reordering, as shown inFIGS. 17A and 17B, is preferable to using either of the frequency domainor the time domain alone due to the strong probability of the directwaves becoming regular. As a result, the effects of the presentinvention are more easily obtained. However, reordering in the frequencydomain may lead to diversity gain due the fact that frequency-domainfluctuations are abrupt. As such, using the frequency and time domainstogether for reordering is not always ideal.

FIGS. 18A and 18B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 17A and 14B. FIG. 18A illustrates afifth example of a frame configuration indicating a reordering schemefor the symbols of modulated signal z1, while FIG. 18B illustrates afifth example of a frame configuration indicating a reordering schemefor the symbols of modulated signal z2. Much like FIGS. 17A and 17B,FIGS. 18A and 18B illustrate the use of the time and frequency domains,together. However, in contrast to FIGS. 17A and 17B, where the frequencydomain is prioritized and the time domain is used for secondary symbolarrangement, FIGS. 18A and 18B prioritize the time domain and use thefrequency domain for secondary symbol arrangement. In FIG. 18B, symbolgroup 1802 corresponds to one period (cycle) of symbols when the phasechanging scheme is used.

In FIGS. 17A, 17B, 18A, and 18B, the reordering scheme applied to thesymbols of modulated signal z1 and the symbols of modulated signal z2may be identical or may differ as in FIGS. 15A and 15B. Both approachesallow good reception quality to be obtained. Also, in FIGS. 17A, 17B,18A, and 18B, the symbols may be arranged non-sequentially as in FIGS.16A and 16B. Both approaches allows good reception quality to beobtained.

FIG. 22 indicates frequency on the horizontal axis and time on thevertical axis thereof, and illustrates an example of a symbol reorderingscheme used by the reorderers 1301A and 1301B from FIG. 13 that differsfrom the above. FIG. 22 illustrates a regular phase changing schemeusing four slots, similar to times u through u+3 from FIG. 6. Thecharacteristic feature of FIG. 22 is that, although the symbols arereordered with respect the frequency domain, when read along the timeaxis, a periodic shift of n (n=1 in the example of FIG. 22) symbols isapparent. The frequency-domain symbol group 2210 in FIG. 22 indicatesfour symbols to which the change of phase is applied at times u throughu+3 from FIG. 6.

Here, symbol #0 is obtained through a change of phase at time u, symbol#1 is obtained through a change of phase at time u+1, symbol #2 isobtained through a change of phase at time u+2, and symbol #3 isobtained through a change of phase at time u+3.

Similarly, for frequency-domain symbol group 2220, symbol #4 is obtainedthrough a change of phase at time u, symbol #5 is obtained through achange of phase at time u+1, symbol #6 is obtained through a change ofphase at time u+2, and symbol #7 is obtained through a change of phaseat time u+3.

The above-described change of phase is applied to the symbol at time $1.However, in order to apply periodic shifting in the time domain, thefollowing phase changes are applied to symbol groups 2201, 2202, 2203,and 2204.

For time-domain symbol group 2201, symbol #0 is obtained through achange of phase at time u, symbol #9 is obtained through a change ofphase at time u+1, symbol #18 is obtained through a change of phase attime u+2, and symbol #27 is obtained through a change of phase at timeu+3.

For time-domain symbol group 2202, symbol #28 is obtained through achange of phase at time u, symbol #1 is obtained through a change ofphase at time u+1, symbol #10 is obtained through a change of phase attime u+2, and symbol #19 is obtained through a change of phase at timeu+3.

For time-domain symbol group 2203, symbol #20 is obtained through achange of phase at time u, symbol #29 is obtained through a change ofphase at time u+1, symbol #2 is obtained through a change of phase attime u+2, and symbol #11 is obtained through a change of phase at timeu+3.

For time-domain symbol group 2204, symbol #12 is obtained through achange of phase at time u, symbol #21 is obtained through a change ofphase at time u+1, symbol #30 is obtained through a change of phase attime u+2, and symbol #3 is obtained through a change of phase at timeu+3.

The characteristic feature of FIG. 22 is seen in that, taking symbol #11as an example, the two neighbouring symbols thereof having the same timein the frequency domain (#10 and #12) are both symbols changed using adifferent phase than symbol #11, and the two neighbouring symbolsthereof having the same carrier in the time domain (#2 and #20) are bothsymbols changed using a different phase than symbol #11. This holds notonly for symbol #11, but also for any symbol having two neighboringsymbols in the frequency domain and the time domain. Accordingly, phasechanging is effectively carried out. This is highly likely to improvedate reception quality as influence from regularizing direct waves isless prone to reception.

Although FIG. 22 illustrates an example in which n=1, the invention isnot limited in this manner. The same may be applied to a case in whichn=3. Furthermore, although FIG. 22 illustrates the realization of theabove-described effects by arranging the symbols in the frequency domainand advancing in the time domain so as to achieve the characteristiceffect of imparting a periodic shift to the symbol arrangement order,the symbols may also be randomly (or regularly) arranged to the sameeffect.

Embodiment 2

In Embodiment 1, described above, phase changing is applied to aweighted (precoded with a fixed precoding matrix) signal z(t). Thefollowing Embodiments describe various phase changing schemes by whichthe effects of Embodiment 1 may be obtained.

In the above-described Embodiment, as shown in FIGS. 3 and 6, phasechanger 317B is configured to perform a change of phase on only one ofthe signals output by the weighting unit 600.

However, phase changing may also be applied before precoding isperformed by the weighting unit 600. In addition to the componentsillustrated in FIG. 6, the transmission device may also feature theweighting unit 600 before the phase changer 317B, as shown in FIG. 25.

In such circumstances, the following configuration is possible. Thephase changer 317B performs a regular change of phase with respect tobaseband signal s2(t), on which mapping has been performed according toa selected modulation scheme, and outputs s2′(t)=s2(t)y(t) (where y(t)varies over time t). The weighting unit 600 executes precoding on s2′t,outputs z2(t)=W2s2′(t) (see Math. 42 (formula 42)) and the result isthen transmitted.

Alternatively, phase changing may be performed on both modulated signalss1(t) and s2(t). As such, the transmission device is configured so as toinclude a phase changer taking both signals output by the weighting unit600, as shown in FIG. 26.

Like phase changer 317B, phase changer 317A performs regular a regularchange of phase on the signal input thereto, and as such changes thephase of signal z1′(t) precoded by the weighting unit. Post-phase changesignal z1(t) is then output to a transmitter.

However, the phase changing rate applied by the phase changers 317A and317B varies simultaneously in order to perform the phase changing shownin FIG. 26. (The following describes a non-limiting example of the phasechanging scheme.) For time u, phase changer 317A from FIG. 26 performsthe change of phase such that z1(t)=y₁(t)z1′(t), while phase changer317B performs the change of phase such that z2(t)=y₂(t)z2′(t). Forexample, as shown in FIG. 26, for time u, y₁(u)=e^(j0) andy₂(u)=e^(−jπ/2), for time u+1, y₁(u+1)=e^(jπ/4) and y₂(u+1)=e^(−j3π/4),and for time u+k, y₁(u+k)=e^(jkπ/4) and y₂(u+k)=e^(j(k3π/4-π/2)). Here,the regular phase changing period (cycle) may be the same for both phasechangers 317A and 317B, or may vary for each.

Also, as described above, a change of phase may be performed beforeprecoding is performed by the weighting unit. In such a case, thetransmission device should be configured as illustrated in FIG. 27.

When a change of phase is carried out on both modulated signals, each ofthe transmit signals is, for example, control information that includesinformation about the phase changing pattern. By obtaining the controlinformation, the reception device knows the phase changing scheme bywhich the transmission device regularly varies the change, i.e., thephase changing pattern, and is thus able to demodulate (decode) thesignals correctly.

Next, variants of the sample configurations shown in FIGS. 6 and 25 aredescribed with reference to FIGS. 28 and 29. FIG. 28 differs from FIG. 6in the inclusion of phase change ON/OFF information 2800 and in that thechange of phase is performed on only one of z1′(t) and z2′(t) (i.e.,performed on one of z1′(t) and z2′(t), which have identical times or acommon frequency). Accordingly, in order to perform the change of phaseon one of z1′(t) and z2′(t), the phase changers 317A and 317B shown inFIG. 28 may each be ON, and performing the change of phase, or OFF, andnot performing the change of phase. The phase change ON/OFF information2800 is control information therefor. The phase change ON/OFFinformation 2800 is output by the signal processing scheme informationgenerator 314 shown in FIG. 3.

Phase changer 317A of FIG. 28 changes the phase to producez1(t)=y₁(t)z1′(t), while phase changer 317B changes the phase to producez2(t)=y₂(t)z2′(t).

Here, a change of phase having a period (cycle) of four is, for example,applied to z1′(t). (Meanwhile, the phase of z2′(t) is not changed.)Accordingly, for time u, y₁(u)=e^(j0) and y₂(u)=1, for time u+1,y₁(u+1)=e^(jπ/2) and y₂(u+1)=1, for time u+2, y₁(u+2)=e^(jπ) andy₂(u+2)=1, and for time u+3, y₁(u+3)=e^(j3π/2) and y₂(u+3)=1.

Next, a change of phase having a period (cycle) of four is, for example,applied to z2′(t). (Meanwhile, the phase of z1′(t) is not changed.)Accordingly, for time u+4, y₁(u+4)=1 and y₂(u+4)=e^(j0), for time u+5,y₁(u+5)=1 and y₂(u+5)=e^(jπ/2), for time u+6, y₁(u+6)=1 andy₂(u+6)=e^(jπ), and for time u+7, y₁(u+7)=1 and y₂(u+7)=e^(j3π/2).

Accordingly, given the above examples.

for any time 8k, y₁(8k)=e^(j0) and y₂(8k)=1,

for any time 8k+1, y₁(8k+1)=e^(jπ/2) and y₂(8k+1)=1,

for any time 8k+2, y₁(8k+2)=e^(jπ) and y₂(8k+2)=1,

for any time 8k+3, y₁(8k+3)=e^(j3π/2) and y₂(8k+3)=1,

for any time 8k+4, y₁(8k+4)=1 and y₂(8k+4)=e^(j0),

for any time 8k+5, y₁(8k+3)=1 and y₂(8k+5)=e^(jπ/2),

for any time 8k+6, y₁(8k+6)=1 and y₂(8k+6)=e^(jπ), and

for any time 8k+7, y₁(8k+7)=1 and y₂(8k+7)=e^(j3π/2).

As described above, there are two intervals, one where the change ofphase is performed on z1′(t) only, and one where the change of phase isperformed on z2′(t) only. Furthermore, the two intervals form a phasechanging period (cycle). While the above explanation describes theinterval where the change of phase is performed on z1′(t) only and theinterval where the change of phase is performed on z2′(t) only as beingequal, no limitation is intended in this manner. The two intervals mayalso differ. In addition, while the above explanation describesperforming a change of phase having a period (cycle) of four on z1′(t)only and then performing a change of phase having a period (cycle) offour on z2′(t) only, no limitation is intended in this manner. Thechanges of phase may be performed on z1′(t) and on z2′(t) in any order(e.g., the change of phase may alternate between being performed onz1′(t) and on z2′(t), or may be performed in random order).

Phase changer 317A of FIG. 29 changes the phase to produces1′(t)=y₁(t)s1(t), while phase changer 317B changes the phase to produces2′(t)=y₂(t)s2(t).

Here, a change of phase having a period (cycle) of four is, for example,applied to s1(t). (Meanwhile, s2(t) remains unchanged). Accordingly, fortime u, y₁(u)=e^(j0) and y₂(u)=1, for time u+1, y₁(u+1)=e^(jπ/2) andy₂(u+1)=1, for time u+2, y₁(u+2)=e^(jπ) and y₂(u+2)=1, and for time u+3,y₁(u+3)=e^(j3π/2) and y₂(u+3)=1.

Next, a change of phase having a period (cycle) of four is, for example,applied to s2(t). (Meanwhile, s1(t) remains unchanged). Accordingly, fortime u+4, y₁(u+4)=1 and y₂(u+4)=e^(j0), for time u+5, y₁(u+5)=1 andy₂(u+5)=e^(jπ/2), for time u+6, y₁(u+6)=1 and y₂(u+6)=e^(jπ), and fortime u+7, y₁(u+7)=1 and y₂(u+7)=e^(j3π/2).

Accordingly, given the above examples,

for any time 8k, y₁(8k)=e^(j0) and y₂(8k)=1,

for any time 8k+1, y₁(8k+1)=e^(jπ/2) and y₂(8k+1)=1,

for any time 8k+2, y₁(8k+2)=e^(jπ) and y₂(8k+2)=1,

for any time 8k+3, y₁(8k+3)=e^(j3π/2) and y₂(8k+3)=1,

for any time 8k+4, y₁(8k+4)=1 and y₂(8k+4)=e^(j0),

for any time 8k+5, y₁(8k+5)=1 and y₂(8k+5)=e^(jπ/2),

for any time 8k+6, y₁(8k+6)=1 and y₂(8k+6)=e^(jπ), and

for any time 8k+7, y₁(8k+7)=1 and y₂(8k+7)=e^(j3π/2).

As described above, there are two intervals, one where the change ofphase is performed on s1(t) only, and one where the change of phase isperformed on s2(t) only. Furthermore, the two intervals form a phasechanging period (cycle). Although the above explanation describes theinterval where the change of phase is performed on s1(t) only and theinterval where the change of phase is performed on s2(t) only as beingequal, no limitation is intended in this manner. The two intervals mayalso differ. In addition, while the above explanation describesperforming the change of phase having a period (cycle) of four on s1(t)only and then performing the change of phase having a period (cycle) offour on s2(t) only, no limitation is intended in this manner. Thechanges of phase may be performed on s1(t) and on s2(t) in any order(e.g., may alternate between being performed on s1(t) and on s2(t), ormay be performed in random order).

Accordingly, the reception conditions under which the reception devicereceives each transmit signal z1 (t) and z2(t) are equalized. Byperiodically switching the phase of the symbols in the received signalsz1(t) and z2(t), the ability of the error corrected codes to correcterrors may be improved, thus ameliorating received signal quality in theLOS environment.

Accordingly, Embodiment 2 as described above is able to produce the sameresults as the previously described Embodiment 1.

Although the present Embodiment used a single-carrier scheme, i.e., timedomain phase changing, as an example, no limitation is intended in thisregard. The same effects are also achievable using multi-carriertransmission. Accordingly, the present Embodiment may also be realizedusing, for example, spread-spectrum communications, OFDM, SC-FDMA(Single Carrier Frequency-Division Multiple Access), SC-OFDM, waveletOFDM as described in Non-Patent Literature 7, and so on. As previouslydescribed, while the present Embodiment explains the change of phase aschanging the phase with respect to the time domain t, the phase mayalternatively be changed with respect to the frequency domain asdescribed in Embodiment 1. That is, considering the phase changingscheme in the time domain t described in the present Embodiment andreplacing t with f (f being the ((sub-)carrier) frequency) leads to achange of phase applicable to the frequency domain. Also, as explainedabove for Embodiment 1, the phase changing scheme of the presentEmbodiment is also applicable to changing the phase with respect boththe time domain and the frequency domain.

Accordingly, although FIGS. 6, 25, 26, and 27 illustrate changes ofphase in the time domain, replacing time t with carrier f in each ofFIGS. 6, 25, 26, and 27 corresponds to a change of phase in thefrequency domain. In other words, replacing (t) with (t, f) where t istime and f is frequency corresponds to performing the change of phase ontime-frequency blocks.

Furthermore, in the present Embodiment, symbols other than data symbols,such as pilot symbols (preamble, unique word, etc) or symbolstransmitting control information, may be arranged within the frame inany manner.

Embodiment 3

Embodiments 1 and 2, described above, discuss regular changes of phase.Embodiment 3 describes a scheme of allowing the reception device toobtain good received signal quality for data, regardless of thereception device arrangement, by considering the location of thereception device with respect to the transmission device.

Embodiment 3 concerns the symbol arrangement within signals obtainedthrough a change of phase.

FIG. 31 illustrates an example of frame configuration for a portion ofthe symbols within a signal in the time-frequency domain, given atransmission scheme where a regular change of phase is performed for amulti-carrier scheme such as OFDM.

First, an example is explained in which the change of phase is performedone of two baseband signals, precoded as explained in Embodiment 1 (seeFIG. 6).

(Although FIG. 6 illustrates a change of phase in the time domain,switching time t with carrier f in FIG. 6 corresponds to a change ofphase in the frequency domain. In other words, replacing (t) with (t,where t is time and f is frequency corresponds to performing phasechanges on time-frequency blocks.)

FIG. 31 illustrates the frame configuration of modulated signal z2′,which is input to phase changer 317B from FIG. 12. Each squarerepresents one symbol (although both signals s1 and s2 are included forprecoding purposes, depending on the precoding matrix, only one ofsignals s1 and s2 may be used).

Consider symbol 3100 at carrier 2 and time $2 of FIG. 31. The carrierhere described may alternatively be termed a sub-carrier.

Within carrier 2, there is a very strong correlation between the channelconditions for symbol 3100 at carrier 2, time $2 and the channelconditions for the time domain nearest-neighbour symbols to time $2,i.e., symbol 3013 at time $1 and symbol 3101 at time $3 within carrier2.

Similarly, for time $2, there is a very strong correlation between thechannel conditions for symbol 3100 at carrier 2, time $2 and the channelconditions for the frequency-domain nearest-neighbour symbols to carrier2, i.e., symbol 3104 at carrier 1, time $2 and symbol 3104 at time $2,carrier 3.

As described above, there is a very strong correlation between thechannel conditions for symbol 3100 and the channel conditions forsymbols 3101, 3102, 3103, and 3104.

The present description considers N different phases (N being aninteger, N≧2) for multiplication in a transmission scheme where thephase is regularly changed. The symbols illustrated in FIG. 31 areindicated as e^(j0), for example. This signifies that this symbol issignal z2′ from FIG. 6 phase-changed through multiplication by e^(j0).That is, the values indicated in FIG. 31 for each of the symbols are thevalues of y(t) from Math. 42 (formula 42), which are also the values ofz2(t)=y₂(t)z2′(t) described in Embodiment 2.

The present Embodiment takes advantage of the high correlation inchannel conditions existing between neighboring symbols in the frequencydomain and/or neighbouring symbols in the time domain in a symbolarrangement enabling high data reception quality to be obtained by thereception device receiving the phase-changed symbols.

In order to achieve this high data reception quality, conditions #1 and#2 are necessary.

(Condition #1)

As shown in FIG. 6, for a transmission scheme involving a regular changeof phase performed on precoded baseband signal z2′ using multi-carriertransmission such as OFDM, time X, carrier Y is a symbol fortransmitting data (hereinafter, data symbol), neighbouring symbols inthe time domain, i.e., at time X−1, carrier Y and at time X+1, carrier Yare also data symbols, and a different change of phase should beperformed on precoded baseband signal z2′ corresponding to each of thesethree data symbols, i.e., on precoded baseband signal z2′ at time X,carrier Y, at time X−1, carrier Y and at time X+1, carrier Y.

(Condition #2)

As shown in FIG. 6, for a transmission scheme involving a regular changeof phase performed on precoded baseband signal z2′ using multi-carriertransmission such as OFDM, time X, carrier Y is a data symbol,neighbouring symbols in the frequency domain, i.e., at time X, carrierY−1 and at time X, carrier Y+1 are also data symbols, and a differentchange of phase should be performed on precoded baseband signal z2′corresponding to each of these three data symbols, i.e., on precodedbaseband signal z2′ at time X, carrier Y, at time X, carrier Y−1 and attime X, carrier Y+1.

Ideally, data symbols satisfying Condition #1 should be present.Similarly, data symbols satisfying Condition #2 should be present.

The reasons supporting Conditions #1 and #2 are as follows.

A very strong correlation exists between the channel conditions of givensymbol of a transmit signal (hereinafter, symbol A) and the channelconditions of the symbols neighbouring symbol A in the time domain, asdescribed above.

Accordingly, when three neighbouring symbols in the time domain eachhave different phases, then despite reception quality degradation in theLOS environment (poor signal quality caused by degradation in conditionsdue to direct wave phase relationships despite high signal quality interms of SNR) for symbol A, the two remaining symbols neighbouringsymbol A are highly likely to provide good reception quality. As aresult, good received signal quality is achievable after errorcorrection and decoding.

Similarly, a very strong correlation exists between the channelconditions of given symbol of a transmit signal (hereinafter, symbol A)and the channel conditions of the symbols neighbouring symbol A in thefrequency domain, as described above.

Accordingly, when three neighbouring symbols in the frequency domaineach have different phases, then despite reception quality degradationin the LOS environment (poor signal quality caused by degradation inconditions due to direct wave phase relationships despite high signalquality in terms of SNR) for symbol A, the two remaining symbolsneighbouring symbol A are highly likely to provide good receptionquality. As a result, good received signal quality is achievable aftererror correction and decoding.

Combining Conditions #1 and #2, ever greater data reception quality islikely achievable for the reception device. Accordingly, the followingCondition #3 can be derived.

(Condition #3)

As shown in FIG. 6, for a transmission scheme involving a regular changeof phase performed on precoded baseband signal z2′ using multi-carriertransmission such as OFDM, time X, carrier Y is a data symbol,neighbouring symbols in the time domain, i.e., at time X−1, carrier Yand at time X+1, carrier Y are also data symbols, and neighbouringsymbols in the frequency domain, i.e., at time X, carrier Y−1 and attime X, carrier Y+1 are also data symbols, and a different change inphase should be performed on precoded baseband signal z2′ correspondingto each of these five data symbols, i.e., on precoded baseband signalz2′ at time X, carrier Y, at time X, carrier Y−1, at time X, carrierY+1, at a time X−1, carrier Y, and at time X+1, carrier Y.

Here, the different changes in phase are as follows. Changes in phaseare defined from 0 radians to 2π radians. For example, for time X,carrier Y, a phase change of e^(jθX,Y) is applied to precoded basebandsignal z2′ from FIG. 6, for time X−1, carrier Y, a phase change ofe^(jθX−1,Y) is applied to precoded baseband signal z2′ from FIG. 6, fortime X+1, carrier Y, a phase change of e^(jθX+1,Y) is applied toprecoded baseband signal z2′ from FIG. 6, such that 0≦θ_(X,Y)≦2π,0≦θ_(X−1,Y)<2π, and 0≦θ_(X+1,Y)<2π, all units being in radians.Accordingly, for Condition #1, it follows that θ_(X,Y)≠θ_(X−1,Y),θ_(X,Y)≠θ_(X+1,Y), and that θ_(X−1,Y)≠θ_(X+1,Y). Similarly, forCondition #2, it follows that θ_(X,Y)≠θ_(X,Y−1), θ_(X,Y)≠θ_(X,Y+1), andthat θ_(X,Y−1)≠θ_(X,Y+1). And, for Condition #3, it follows thatθ_(X,Y)≠θ_(X−1,Y), θ_(X,Y)≠θ_(X+1,Y), θ_(X,Y)≠θ_(X,Y−1),θ_(X,Y)≠θ_(X,Y−1), θ_(X−1,Y)≠θ_(X+1,Y)≠θ_(X,Y−1), θ_(X−1,Y)≠θ_(X+1,Y),θ_(X+1,Y)≠θ_(X+1,Y), θ_(X+1,Y)≠θ_(X,Y+1), and that θ_(X,Y−1)≠θ_(X,Y+1).

Ideally, a data symbol should satisfy Condition #3.

FIG. 31 illustrates an example of Condition #3 where symbol Acorresponds to symbol 3100. The symbols are arranged such that the phaseby which precoded baseband signal z2′ from FIG. 6 is multiplied differsfor symbol 3100, for both neighbouring symbols thereof in the timedomain 3101 and 3102, and for both neighbouring symbols thereof in thefrequency domain 3102 and 3104. Accordingly, despite received signalquality degradation of symbol 3100 for the receiver, good signal qualityis highly likely for the neighbouring signals, thus guaranteeing goodsignal quality after error correction.

FIG. 32 illustrates a symbol arrangement obtained through phase changesunder these conditions.

As evident from FIG. 32, with respect to any data symbol, a differentchange in phase is applied to each neighbouring symbol in the timedomain and in the frequency domain. As such, the ability of thereception device to correct errors may be improved.

In other words, in FIG. 32, when all neighbouring symbols in the timedomain are data symbols, Condition #1 is satisfied for all Xs and allYs.

Similarly, in FIG. 32, when all neighbouring symbols in the frequencydomain are data symbols, Condition #2 is satisfied for all Xs and allYs.

Similarly, in FIG. 32, when all neighbouring symbols in the frequencydomain are data symbols and all neighbouring symbols in the time domainare data symbols, Condition #3 is satisfied for all Xs and all Ys.

The following describes an example in which a change of phase isperformed on two precoded baseband signals, as explained in Embodiment 2(see FIG. 26).

When a change of phase is performed on precoded baseband signal z1′ andprecoded baseband signal z2′ as shown in FIG. 26, several phase changingschemes are possible. The details thereof are explained below.

Scheme 1 involves a change in phase performed on precoded basebandsignal z2′ as described above, to achieve the change in phaseillustrated by FIG. 32. In FIG. 32, a change of phase having a period(cycle) of 10 is applied to precoded baseband signal z2′. However, asdescribed above, in order to satisfy Conditions #1, #2, and #3, thechange in phase applied to precoded baseband signal z2′ at each(sub-)carrier varies over time. (Although such changes are applied inFIG. 32 with a period (cycle) of ten, other phase changing schemes arealso possible.) Then, as shown in FIG. 33, the change in phase performedon precoded baseband signal z1′ produces a constant value that isone-tenth of that of the change in phase performed on precoded basebandsignal z2′. In FIG. 33, for a period (cycle) (of change in phaseperformed on precoded baseband signal z2′) including time $1, the valueof the change in phase performed on precoded baseband signal z1′ ise^(j0). Then, for the next period (cycle) (of change in phase performedon precoded baseband signal z2′) including time $2, the value of thechange in phase performed on precoded baseband signal z1′ is e^(jπ/9),and so on.

The symbols illustrated in FIG. 33 are indicated as e^(j0), for example.This signifies that this symbol is signal z1′ from FIG. 26 on which achange in phase as been applied through multiplication by e^(j0). Thatis, the values indicated in FIG. 33 for each of the symbols are thevalues of z1′(t)=y₂(t)z1′(t) described in Embodiment 2 for y₁(t).

As shown in FIG. 33, the change in phase performed on precoded basebandsignal z1′ produces a constant value that is one-tenth that of thechange in phase performed on precoded baseband signal z2′ such that thepost-phase change value varies with the number of each period (cycle).(As described above, in FIG. 33, the value is e^(j0) for the firstperiod (cycle), e^(jπ/9) for the second period (cycle), and so on.)

As described above, the change in phase performed on precoded basebandsignal z2′ has a period (cycle) of ten, but the period (cycle) can beeffectively made greater than ten by taking the change in phase appliedto precoded baseband signal z1′ and to precoded baseband signal z2′ intoconsideration. Accordingly, data reception quality may be improved forthe reception device.

Scheme 2 involves a change in phase of precoded baseband signal z2′ asdescribed above, to achieve the change in phase illustrated by FIG. 32.In FIG. 32, a change of phase having a period (cycle) of ten is appliedto precoded baseband signal z2′. However, as described above, in orderto satisfy Conditions #1, #2, and #3, the change in phase applied toprecoded baseband signal z2′ at each (sub-)carrier varies over time.(Although such changes are applied in FIG. 32 with a period (cycle) often, other phase changing schemes are also possible.) Then, as shown inFIG. 30, the change in phase performed on precoded baseband signal z1′differs from that performed on precoded baseband signal z2′ in having aperiod (cycle) of three rather than ten.

The symbols illustrated in FIG. 30 are indicated as e^(j0), for example.This signifies that this symbol is signal z1′ from FIG. 26 to which achange in phase has been applied through multiplication by e^(j0). Thatis, the values indicated in FIG. 30 for each of the symbols are thevalues of z1(t)=y₁(t)z1′(t) described in Embodiment 2 for y₁(t).

As described above, the change in phase performed on precoded basebandsignal z2′ has a period (cycle) of ten, but by taking the changes inphase applied to precoded baseband signal z1′ and precoded basebandsignal z2′ into consideration, the period (cycle) can be effectivelymade equivalent to 30 for both precoded baseband signals z1′ and z2′.Accordingly, data reception quality may be improved for the receptiondevice. An effective way of applying scheme 2 is to perform a change inphase on precoded baseband signal z1′ with a period (cycle) of N andperform a change in phase on precoded baseband signal z2′ with a period(cycle) of M such that N and M are coprime. As such, by taking bothprecoded baseband signals z1′ and z2′ into consideration, a period(cycle) of N×M is easily achievable, effectively making the period(cycle) greater when N and M are coprime.

The above describes an example of the phase changing scheme pertainingto Embodiment 3. The present invention is not limited in this manner. Asexplained for Embodiments 1 and 2, a change in phase may be performedwith respect the frequency domain or the time domain, or ontime-frequency blocks. Similar improvement to the data reception qualitycan be obtained for the reception device in all cases.

The same also applies to frames having a configuration other than thatdescribed above, where pilot symbols (SP (Scattered Pilot) and symbolstransmitting control information are inserted among the data symbols.The details of change in phase in such circumstances are as follows.

FIGS. 47A and 47B illustrate the frame configuration of modulatedsignals (precoded baseband signals) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 47A illustrates the frame configuration ofmodulated signal (precoded baseband signals) z1 or z1′ while FIG. 47Billustrates the frame configuration of modulated signal (precodedbaseband signals) z2′. In FIGS. 47A and 47B, 4701 marks pilot symbolswhile 4702 marks data symbols. The data symbols 4702 are symbols onwhich precoding or precoding and a change in phase have been performed.

FIGS. 47A and 47B, like FIG. 6, indicate the arrangement of symbols whena change in phase is applied to precoded baseband signal z2′ (while nochange of phase is performed on precoded baseband signal z1). (AlthoughFIG. 6 illustrates a change in phase with respect to the time domain,switching time t with carrier fin FIG. 6 corresponds to a change inphase with respect to the frequency domain. In other words, replacing(t) with (t, f) where t is time and f is frequency corresponds toperforming a change of phase on time-frequency blocks.) Accordingly, thenumerical values indicated in FIGS. 47A and 47B for each of the symbolsare the values of precoded baseband signal z2′ after the change inphase. No values are given for the symbols of precoded baseband signalz1′ (z1) as no change in phase is performed thereon.

The key point of FIGS. 47A and 47B is that the change in phase isperformed on the data symbols of precoded baseband signal z2′, i.e., onprecoded symbols. (The symbols under discussion, being precoded,actually include both symbols s1 and s2.) Accordingly, no change ofphase is performed on the pilot symbols inserted into z2′.

FIGS. 48A and 48B illustrate the frame configuration of modulatedsignals (precoded baseband signals) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 48A illustrates the frame configuration ofmodulated signal (precoded baseband signals) z1 or z1′ while FIG. 47Billustrates the frame configuration of modulated signal (precodedbaseband signals) z2′. In FIGS. 48A and 48B, 4701 marks pilot symbolswhile 4702 marks data symbols. The data symbols 4702 are symbols onwhich precoding, or precoding and a change in phase, have beenperformed.

FIGS. 48A and 48B, like FIG. 26, indicate the arrangement of symbolswhen a change in phase is applied to precoded baseband signal z1′ and toprecoded baseband signal z2′. (Although FIG. 26 illustrates a change inphase with respect to the time domain, switching time t with carrier fin FIG. 26 corresponds to a change in phase with respect to thefrequency domain. In other words, replacing (t) with (t, f) where t istime and f is frequency corresponds to performing a change of phase ontime-frequency blocks.) Accordingly, the numerical values indicated inFIGS. 48A and 48B for each of the symbols are the values of precodedbaseband signal z1′ and z2′ after the change in phase.

The key point of FIG. 47 is that a change of phase is performed on thedata symbols of precoded baseband signal z1′, that is, on the precodedsymbols thereof, and on the data symbols of precoded baseband signalz2′, that is, on the precoded symbols thereof. (The symbols underdiscussion, being precoded, actually include both symbols s1 and s2.)Accordingly, no change of phase is performed on the pilot symbolsinserted in z1′, nor on the pilot symbols inserted in z2′.

FIGS. 49A and 49B illustrate the frame configuration of modulatedsignals (precoded baseband signals) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 49A illustrates the frame configuration ofmodulated signal (precoded baseband signals) z1 or z1′ while FIG. 49Billustrates the frame configuration of modulated signal (precodedbaseband signal) z2′. In FIGS. 49A and 49B, 4701 marks pilot symbols,4702 marks data symbols, and 4901 marks null symbols for which thein-phase component of the baseband signal I=0 and the quadraturecomponent Q=0. As such, data symbols 4702 are symbols on which precodingor precoding and the change in phase have been performed. FIGS. 49A and49B differ from FIGS. 47A and 47B in the configuration scheme forsymbols other than data symbols. The times and carriers at which pilotsymbols are inserted into modulated signal z1′ are null symbols inmodulated signal z2′. Conversely, the times and carriers at which pilotsymbols are inserted into modulated signal z2′ are null symbols inmodulated signal z1′.

FIGS. 49A and 49B, like FIG. 6, indicate the arrangement of symbols whena change in phase is applied to precoded baseband signal z2′ (while nochange of phase is performed on precoded baseband signal z1). (AlthoughFIG. 6 illustrates a change of phase with respect to the time domain,switching time t with carrier fin FIG. 6 corresponds to a change ofphase with respect to the frequency domain. In other words, replacing(t) with (t, f) where t is time and f is frequency corresponds toperforming a change of phase on time-frequency blocks.) Accordingly, thenumerical values indicated in FIGS. 49A and 49B for each of the symbolsare the values of precoded baseband signal z2′ after a change of phaseis performed. No values are given for the symbols of precoded basebandsignal z1′ (z1) as no change of phase is performed thereon.

The key point of FIGS. 49A and 49B is that a change of phase isperformed on the data symbols of precoded baseband signal z2′, i.e., onprecoded symbols. (The symbols under discussion, being precoded,actually include both symbols s1 and s2.) Accordingly, no change ofphase is performed on the pilot symbols inserted into z2′.

FIGS. 50A and 50B illustrate the frame configuration of modulatedsignals (precoded baseband signals) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 50A illustrates the frame configuration ofmodulated signal (precoded baseband signal) z1 or z1′ while FIG. 50Billustrates the frame configuration of modulated signal (precodedbaseband signal) z2′. In FIGS. 50A and 50B, 4701 marks pilot symbols,4702 marks data symbols, and 4901 marks null symbols for which thein-phase component of the baseband signal I=0 and the quadraturecomponent Q=0. As such, data symbols 4702 are symbols on whichprecoding, or precoding and a change of phase, have been performed.FIGS. 50A and 50B differ from FIGS. 48A and 48B in the configurationscheme for symbols other than data symbols. The times and carriers atwhich pilot symbols are inserted into modulated signal z1′ are nullsymbols in modulated signal z2′. Conversely, the times and carriers atwhich pilot symbols are inserted into modulated signal z2′ are nullsymbols in modulated signal z1′.

FIGS. 50A and 50B, like FIG. 26, indicate the arrangement of symbolswhen a change of phase is applied to precoded baseband signal z1′ and toprecoded baseband signal z2′. (Although FIG. 26 illustrates a change ofphase with respect to the time domain, switching time t with carrier fin FIG. 26 corresponds to a change of phase with respect to thefrequency domain. In other words, replacing (t) with (t, f) where t istime and f is frequency corresponds to performing a change of phase ontime-frequency blocks.) Accordingly, the numerical values indicated inFIGS. 50A and 50B for each of the symbols are the values of precodedbaseband signal z1′ and z2′ after a change of phase.

The key point of FIGS. 50A and 50B is that a change of phase isperformed on the data symbols of precoded baseband signal z1′, that is,on the precoded symbols thereof, and on the data symbols of precodedbaseband signal z2′, that is, on the precoded symbols thereof. (Thesymbols under discussion, being precoded, actually include both symbolss1 and s2.) Accordingly, no change of phase is performed on the pilotsymbols inserted in z1′, nor on the pilot symbols inserted in z2′.

FIG. 51 illustrates a sample configuration of a transmission devicegenerating and transmitting modulated signal having the frameconfiguration of FIGS. 47A, 47B, 49A, and 49B. Components thereofperforming the same operations as those of FIG. 4 use the same referencesymbols there as.

In FIG. 51, the weighting units 308A and 308B and phase changer 317Bonly operate at times indicated by the frame configuration signal 313 ascorresponding to data symbols.

In FIG. 51, a pilot symbol generator 5101 (that also generates nullsymbols) outputs baseband signals 5102A and 5102B for a pilot symbolwhenever the frame configuration signal 313 indicates a pilot symbol (ora null symbol).

Although not indicated in the frame configurations from FIGS. 47Athrough 50B, when precoding (or phase rotation) is not performed, suchas when transmitting a modulated signal using only one antenna (suchthat the other antenna transmits no signal) or when using a space-timecoding transmission scheme (particularly, space-time block coding) totransmit control information symbols, then the frame configurationsignal 313 takes control information symbols 5104 and controlinformation 5103 as input. When the frame configuration signal 313indicates a control information symbol, baseband signals 5102A and 5102Bthereof are output.

Wireless units 310A and 310B of FIG. 51 take a plurality of basebandsignals as input and select a desired baseband signal according to theframe configuration signal 313. Wireless units 310A and 310B then applyOFDM signal processing and output modulated signals 311A and 311Bconforming to the frame configuration.

FIG. 52 illustrates a sample configuration of a transmission devicegenerating and transmitting modulated signal having the frameconfiguration of FIGS. 48A, 48B, 50A, and 50B. Components thereofperforming the same operations as those of FIGS. 4 and 51 use the samereference symbols there as. FIG. 51 features an additional phase changer317A that only operates when the frame configuration signal 313indicates a data symbol. At all other times, the operations areidentical to those explained for FIG. 51.

FIG. 53 illustrates a sample configuration of a transmission device thatdiffers from that of FIG. 51. The following describes the points ofdifference. As shown in FIG. 53, phase changer 317B takes a plurality ofbaseband signals as input. Then, when the frame configuration signal 313indicates a data symbol, phase changer 317B performs a change of phaseon precoded baseband signal 316B. When frame configuration signal 313indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 317B pauses phase changing operations, such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

A selector 5301 takes the plurality of baseband signals as input andselects a baseband signal having a symbol indicated by the frameconfiguration signal 313 for output.

FIG. 54 illustrates a sample configuration of a transmission device thatdiffers from that of FIG. 52. The following describes the points ofdifference. As shown in FIG. 54, phase changer 317B takes a plurality ofbaseband signals as input. Then, when the frame configuration signal 313indicates a data symbol, phase changer 317B performs a change of phaseon precoded baseband signal 316B. When frame configuration signal 313indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 317B pauses phase changing operations such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

Similarly, as shown in FIG. 54, phase changer 5201 takes a plurality ofbaseband signals as input. Then, when the frame configuration signal 313indicates a data symbol, phase changer 5201 performs a change of phaseon precoded baseband signal 309A. When frame configuration signal 313indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 5201 pauses phase changing operations such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

The above explanations are given using pilot symbols, control symbols,and data symbols as examples. However, the present invention is notlimited in this manner. When symbols are transmitted using schemes otherthan precoding, such as single-antenna transmission or transmissionusing space-time block coding, not performing a change of phase isimportant. Conversely, performing a change of phase on symbols that havebeen precoded is the key point of the present invention.

Accordingly, a characteristic feature of the present invention is thatthe change of phase is not performed on all symbols within the frameconfiguration in the time-frequency domain, but only performed onsignals that have been precoded.

Embodiment 4

Embodiments 1 and 2, described above, discuss a regular change of phase.Embodiment 3, however, discloses performing a different change of phaseon neighbouring symbols.

The present Embodiment describes a phase changing scheme that variesaccording to the modulation scheme and the coding rate of theerror-correcting codes used by the transmission device.

Table 1, below, is a list of phase changing scheme settingscorresponding to the settings and parameters of the transmission device.

TABLE 1 No. of Modulated Phase Transmission Changing Signals ModulationScheme Coding Rate Pattern 2 #1: QPSK, #2: QPSK #1: 1/2, #2 2/3 #1: —,#2: A 2 #1: QPSK, #2: QPSK #1: 1/2, #2: 3/4 #1: A, #2: B 2 #1: QPSK, #2:QPSK #1: 2/3, #2: 3/5 #1: A, #2: C 2 #1: QPSK, #2: QPSK #1: 2/3, #2: 2/3#1: C, #2: — 2 #1: QPSK, #2: QPSK #1: 3/3, #2: 2/3 #1: D, #2: E 2 #1:QPSK, #2: 16-QAM #1: 1/2, #2: 2/3 #1: B, #2: A 2 #1: QPSK, #2: 16-QAM#1: 1/2, #2: 3/4 #1: A, #2: C 2 #1: QPSK, #2: 16-QAM #1: 1/2, #2: 3/5#1: —, #2: E 2 #1: QPSK, #2: 16-QAM #1: 2/3, #2: 3/4 #1: D, #2: — 2 #1:QPSK, #2: 16-QAM #1: 2/3, #2: 5/6 #1: D, #2: B 2 #1: 16-QAM, #2: 16-QAM#1: 1/2, #2: 2/3 #1: —, #2: E . . . . . . . . . . . .

In Table 1, #1 denotes modulated signal s1 from Embodiment 1 describedabove (baseband signal s1 modulated with the modulation scheme set bythe transmission device) and #2 denotes modulated signal s2 (basebandsignal s2 modulated with the modulation scheme set by the transmissiondevice). The coding rate column of Table 1 indicates the coding rate ofthe error-correcting codes for modulation schemes #1 and #2. The phasechanging pattern column of Table 1 indicates the phase changing schemeapplied to precoded baseband signals z1 (z1′) and z2 (z2′), as explainedin Embodiments 1 through 3. Although the phase changing patterns arelabeled A, B, C, D, E, and so on, this refers to the phase change degreeapplied, for example, in a phase changing pattern given by Math. 46(formula 46) and Math. 47 (formula 47), above. In the phase changingpattern column of Table 1, the dash signifies that no change of phase isapplied.

The combinations of modulation scheme and coding rate listed in Table 1are examples. Other modulation schemes (such as 128-QAM and 256-QAM) andcoding rates (such as 7/8) not listed in Table 1 may also be included.Also, as described in Embodiment 1, the error-correcting codes used fors1 and s2 may differ (Table 1 is given for cases where a single type oferror-correcting codes is used, as in FIG. 4). Furthermore, the samemodulation scheme and coding rate may be used with different phasechanging patterns. The transmission device transmits informationindicating the phase changing patterns to the reception device. Thereception device specifies the phase changing pattern bycross-referencing the information and Table 1, then performsdemodulation and decoding. When the modulation scheme anderror-correction scheme determine a unique phase changing pattern, thenas long as the transmission device transmits the modulation scheme andinformation regarding the error-correction scheme, the reception deviceknows the phase changing pattern by obtaining that information. As such,information pertaining to the phase changing pattern is not strictlynecessary.

In Embodiments 1 through 3, the change of phase is applied to precodedbaseband signals. However, the amplitude may also be modified along withthe phase in order to apply periodical, regular changes. Accordingly, anamplification modification pattern regularly modifying the amplitude ofthe modulated signals may also be made to conform to Table 1. In suchcircumstances, the transmission device should include an amplificationmodifier that modifies the amplification after weighting unit 308A orweighting unit 308B from FIG. 3 or 4. In addition, amplificationmodification may be performed on only one of or on both of the precodedbaseband signals z1(t) and z2(t) (in the former case, the amplificationmodifier is only needed after one of weighting unit 308A and 308B).

Furthermore, although not indicated in Table 1 above, the mapping schememay also be regularly modified by the mapper, without a regular changeof phase.

That is, when the mapping scheme for modulated signal s1(t) is 16-QAMand the mapping scheme for modulated signal s2(t) is also 16-QAM, themapping scheme applied to modulated signal s2(t) may be regularlychanged as follows: from 16-QAM to 16-APSK, to 16-QAM in the IQ plane,to a first mapping scheme producing a signal point layout unlike16-APSK, to 16-QAM in the IQ plane, to a second mapping scheme producinga signal point layout unlike 16-APSK, and so on. As such, the datareception quality can be improved for the reception device, much likethe results obtained by a regular change of phase described above.

In addition, the present invention may use any combination of schemesfor a regular change of phase, mapping scheme, and amplitude, and thetransmit signal may transmit with all of these taken into consideration.

The present Embodiment may be realized using single-carrier schemes aswell as multi-carrier schemes. Accordingly, the present Embodiment mayalso be realized using, for example, spread-spectrum communications,OFDM, SC-FDM, SC-OFDM, wavelet OFDM as described in Non-PatentLiterature 7, and so on. As described above, the present Embodimentdescribes changing the phase, amplitude, and mapping schemes byperforming phase, amplitude, and mapping scheme modifications withrespect to the time domain t. However, much like Embodiment 1, the samechanges may be carried out with respect to the frequency domain. Thatis, considering the phase, amplitude, and mapping scheme modification inthe time domain t described in the present Embodiment and replacing twith f (f being the ((sub-) carrier) frequency) leads to phase,amplitude, and mapping scheme modification applicable to the frequencydomain. Also, the phase, amplitude, and mapping scheme modification ofthe present Embodiment is also applicable to phase, amplitude, andmapping scheme modification in both the time domain and the frequencydomain.

Furthermore, in the present Embodiment, symbols other than data symbols,such as pilot symbols (preamble, unique word, etc) or symbolstransmitting control information, may be arranged within the frame inany manner.

Embodiment A1

The present Embodiment describes a scheme for regularly changing thephase when encoding is performed using block codes as described inNon-Patent Literature 12 through 15, such as QC (Quasi-Cyclic) LDPCCodes (not only QC-LDPC but also LDPC codes may be used), concatenatedLDPC and BCH (Bose-Chaudhuri-Hocquenghem) codes, Turbo codes orDuo-Binary Turbo Codes using tail-biting, and so on. The followingexample considers a case where two streams s1 and s2 are transmitted.However, when encoding has been performed using block codes and controlinformation and the like is not required, the number of bits making upeach coded block matches the number of bits making up each block code(control information and so on described below may yet be included).When encoding has been performed using block codes or the like andcontrol information or the like (e.g., CRC (cyclic redundancy check)transmission parameters) is required, then the number of bits making upeach coded block is the sum of the number of bits making up the blockcodes and the number of bits making up the information.

FIG. 34 illustrates the varying numbers of symbols and slots needed ineach coded block when block codes are used. FIG. 34 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 4, and thetransmission device has only one encoder. (Here, the transmission schememay be any single-carrier scheme or multi-carrier scheme such as OFDM.)

As shown in FIG. 34, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for64-QAM.

Then, given that the transmission device from FIG. 4 transmits twostreams simultaneously, 1500 of the aforementioned 3000 symbols neededwhen the modulation scheme is QPSK are assigned to s1 and the other 1500symbols are assigned to s2. As such, 1500 slots for transmitting the1500 symbols (hereinafter, slots) are required for each of s1 and s2.

By the same reasoning, when the modulation scheme is 16-QAM, 750 slotsare needed to transmit all of the bits making up a single coded block,and when the modulation scheme is 64-QAM, 500 slots are needed totransmit all of the bits making up a single coded block.

The following describes the relationship between the above-defined slotsand the phase of multiplication, as pertains to schemes for a regularchange of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase. That is, five different phase changing values (or phasechanging sets) have been prepared for the phase changer of thetransmission device from FIG. 4 (equivalent to the period (cycle) fromEmbodiments 1 through 4) (As in FIG. 6, five phase changing values areneeded in order to perform a change of phase with a period (cycle) offive on precoded baseband signal z2′ only. Also, as in FIG. 26, twophase changing values are needed for each slot in order to perform thechange of phase on both precoded baseband signals z1′ and z2′. These twophase changing values are termed a phase changing set. Accordingly, fivephase changing sets should ideally be prepared in order to perform thechange of phase with a period (cycle) of five in such circumstances).These five phase changing values (or phase changing sets) are expressedas PHASE[0], PHASE[1], PHASE[2], PHASE[3], and PHASE[4].

For the above-described 1500 slots needed to transmit the 6000 bitsmaking up a single coded block when the modulation scheme is QPSK,PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots, PHASE[2]is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] isused on 300 slots. This is due to the fact that any bias in phase usagecauses great influence to be exerted by the more frequently used phase,and that the reception device is dependent on such influence for datareception quality.

Similarly, for the above-described 700 slots needed to transmit the 6000bits making up a single coded block when the modulation scheme is16-QAM, PHASE[0] is used on 150 slots, PHASE[1] is used on 150 slots,PHASE[2] is used on 150 slots, PHASE[3] is used on 150 slots, andPHASE[4] is used on 150 slots.

Furthermore, for the above-described 500 slots needed to transmit the6000 bits making up a single coded block when the modulation scheme is64-QAM, PHASE[0] is used on 100 slots, PHASE[1] is used on 100 slots,PHASE[2] is used on 100 slots, PHASE[3] is used on 100 slots, andPHASE[4] is used on 100 slots.

As described above, a scheme for a regular change of phase requires thepreparation of N phase changing values (or phase changing sets) (wherethe N different phases are expressed as PHASE[0], PHASE[1], PHASE[2] . .. PHASE[N−2], PHASE[N−1]). As such, in order to transmit all of the bitsmaking up a single coded block, PHASE[0] is used on K₀ slots, PHASE[1]is used on K₁ slots, PHASE[i] is used on K_(i) slots (where i=0, 1, 2 .. . N−1 (i denotes an integer that satisfies 0≦i≦N−1)), and PHASE[N−1]is used on K_(N−1) slots, such that Condition #A01 is met.

(Condition #A01)

K₀=K₁ . . . =K_(i)=K_(N−1). That is, K_(a)=K_(b) (∀a and ∀b where a, b,=0, 1, 2 . . . N−1 (a denotes an integer that satisfies 0≦a≦N−1, bdenotes an integer that satisfies 0≦b≦N−1), a≠b).

Then, when a communication system that supports multiple modulationschemes selects one such supported modulation scheme for use, Condition#A01 is preferably satisfied for the supported modulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbols (though some may happen to use the same number),Condition #A01 may not be satisfied for some modulation schemes. In sucha case, the following condition applies instead of Condition #A01.

(Condition #A02)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≦a≦N−1, b denotes an integer thatsatisfies 0≦b≦N−1), a≠b)

FIG. 35 illustrates the varying numbers of symbols and slots needed intwo coded blocks when block codes are used. FIG. 35 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 3 and FIG.12, and the transmission device has two encoders. (Here, thetransmission scheme may be any single-carrier scheme or multi-carrierscheme such as OFDM.)

As shown in FIG. 35, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for64-QAM.

The transmission device from FIG. 3 and the transmission device fromFIG. 12 each transmit two streams at once, and have two encoders. Assuch, the two streams each transmit different code blocks. Accordingly,when the modulation scheme is QPSK, two coded blocks drawn from s1 ands2 are transmitted within the same interval, e.g., a first coded blockdrawn from s1 is transmitted, then a second coded block drawn from s2 istransmitted. As such, 3000 slots are needed in order to transmit thefirst and second coded blocks.

By the same reasoning, when the modulation scheme is 16-QAM, 1500 slotsare needed to transmit all of the bits making up the two coded blocks,and when the modulation scheme is 64-QAM, 1000 slots are needed totransmit all of the bits making up the two coded blocks.

The following describes the relationship between the above-defined slotsand the phase of multiplication, as pertains to schemes for a regularchange of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase. That is, five different phase changing values (or phasechanging sets) have been prepared for the phase changers of thetransmission devices from FIGS. 3 and 12 (equivalent to the period(cycle) from Embodiments 1 through 4) (As in FIG. 6, five phase changingvalues are needed in order to perform a change of phase having a period(cycle) of five on precoded baseband signal z2′ only. Also, as in FIG.26, two phase changing values are needed for each slot in order toperform the change of phase on both precoded baseband signals z1′ andz2′. These two phase changing values are termed a phase changing set.Accordingly, five phase changing sets should ideally be prepared inorder to perform the change of phase with a period (cycle) of five insuch circumstances). These five phase changing values (or phase changingsets) are expressed as PHASE[0], PHASE[1], PHASE[2], PHASE[3], andPHASE[4].

For the above-described 3000 slots needed to transmit the 6000×2 bitsmaking up a single coded block when the modulation scheme is QPSK,PHASE[0] is used on 600 slots, PHASE[1] is used on 600 slots, PHASE[2]is used on 600 slots, PHASE[3] is used on 600 slots, and PHASE[4] isused on 600 slots. This is due to the fact that any bias in phase usagecauses great influence to be exerted by the more frequently used phase,and that the reception device is dependent on such influence for datareception quality.

Furthermore, in order to transmit the first coded block, PHASE[0] isused on slots 600 times, PHASE[1] is used on slots 600 times, PHASE[2]is used on slots 600 times, PHASE[3] is used on slots 600 times, andPHASE[4] is used on slots 600 times. Furthermore, in order to transmitthe second coded block, PHASE[0] is used on slots 600 times, PHASE[1] isused on slots 600 times, PHASE[2] is used on slots 600 times, PHASE[3]is used on slots 600 times, and PHASE[4] is used on slots 600 times.

Similarly, for the above-described 1500 slots needed to transmit the6000×2 bits making up the two coded blocks when the modulation scheme is16-QAM, PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots,PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, andPHASE[4] is used on 300 slots.

Furthermore, in order to transmit the first coded block, PHASE[0] isused on slots 300 times, PHASE[1] is used on slots 300 times, PHASE[2]is used on slots 300 times, PHASE[3] is used on slots 300 times, andPHASE[4] is used on slots 300 times. Furthermore, in order to transmitthe second coded block, PHASE[0] is used on slots 300 times, PHASE[1] isused on slots 300 times, PHASE[2] is used on slots 300 times, PHASE[3]is used on slots 300 times, and PHASE[4] is used on slots 300 times.

Similarly, for the above-described 1000 slots needed to transmit the6000×2 bits making up the two coded blocks when the modulation scheme is64-QAM, PHASE[0] is used on 200 slots, PHASE[1] is used on 200 slots,PHASE[2] is used on 200 slots, PHASE[3] is used on 200 slots, andPHASE[4] is used on 200 slots.

Furthermore, in order to transmit the first coded block, PHASE[0] isused on slots 200 times, PHASE[1] is used on slots 200 times, PHASE[2]is used on slots 200 times, PHASE[3] is used on slots 200 times, andPHASE[4] is used on slots 200 times. Furthermore, in order to transmitthe second coded block, PHASE[0] is used on slots 200 times, PHASE[1] isused on slots 200 times, PHASE[2] is used on slots 200 times, PHASE[3]is used on slots 200 times, and PHASE[4] is used on slots 200 times.

As described above, a scheme for regularly changing the phase requiresthe preparation of phase changing values (or phase changing sets)expressed as PHASE[0], PHASE[1], PHASE[2] . . . PHASE[N−2], PHASE[N−1].As such, in order to transmit all of the bits making up two codedblocks, PHASE[0] is used on K_(o) slots, PHASE[1] is used on K₁ slots,PHASE[i] is used on K_(i) slots (where i=0, 1, 2 . . . N−1 (i denotes aninteger that satisfies 0≦i≦N−1), and PHASE[N−1] is used on K_(N−1)slots, such that Condition #A03 is met.

(Condition #A03)

K₀=K₁ . . . =K_(i)= . . . K_(N−1). That is, K_(a)=K_(b) (∀a and ∀b wherea, b, =0, 1, 2 . . . N−1 (a denotes an integer that satisfies 0≦a≦N−1, bdenotes an integer that satisfies 0≦b≦N−1), a≠b).

Further, in order to transmit all of the bits making up the first codedblock, PHASE[0] is used K_(0,1) times, PHASE[1] is used K_(1,1) times,PHASE[i] is used K_(i,1) times (where i=0, 1, 2 . . . N−1 (i denotes aninteger that satisfies 0≦i≦N−1), and PHASE[N−1] is used K_(N−1,1) times,such that Condition #A04 is met.

(Condition #A04)

K_(0,1)=K_(1,1)= . . . K_(i,1)= . . . K_(N−1,1). That is,K_(a,1)=K_(b,1) (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a denotes aninteger that satisfies 0≦a≦N−1, b denotes an integer that satisfies0≦b≦N−1), a≠b).Furthermore, in order to transmit all of the bits making up the secondcoded block, PHASE[0] is used K_(0,2) times, PHASE[1] is used K_(1,2)times, PHASE[i] is used K_(i,2) times (where i=0, 1, 2 . . . N−1 (idenotes an integer that satisfies 0≦i≦N−1), and PHASE[N−1] is usedK_(N−1,2) times, such that Condition #A05 is met.

(Condition #A05)

K_(0,2)=K_(1,2)= . . . K_(i,2)= . . . K_(N−1,2). That is,K_(a,2)=K_(b,2) (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a denotes aninteger that satisfies 0≦a≦N−1, b denotes an integer that satisfies0≦b≦N−1), a≠b).

Then, when a communication system that supports multiple modulationschemes selects one such supported modulation scheme for use, Condition#A03, #A04, and #A05 should preferably be met for the supportedmodulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbol (though some may happen to use the same number),Conditions #A03, #A04, and #A05 may not be satisfied for some modulationschemes. In such a case, the following conditions apply instead ofCondition #A03, #A04, and #A05.

(Condition #A06)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≦a≦N−1, b denotes an integer thatsatisfies 0≦b≦N−1), a≠b).

(Condition #A07)

The difference between K_(a,1) and K_(b,1) satisfies 0 or 1. That is,|K_(a,1)−K_(b,1)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . .N−1, (a denotes an integer that satisfies 0≦a≦N−1, b denotes an integerthat satisfies 0≦b≦N−1) a≠b).

(Condition #A08)

The difference between K_(a,2) and K_(b,2) satisfies 0 or 1. That is,|K_(a,2)−Kb,2| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≦a≦N−1, b denotes an integer thatsatisfies 0≦b≦N−1), a≠b).

As described above, bias among the phases being used to transmit thecoded blocks is removed by creating a relationship between the codedblock and the phase of multiplication. As such, data reception qualitycan be improved for the reception device.

In the present Embodiment N phase changing values (or phase changingsets) are needed in order to perform a change of phase having a period(cycle) of N with the scheme for a regular change of phase. As such, Nphase changing values (or phase changing sets) PHASE[0], PHASE[1],PHASE[2] . . . PHASE[N−2], and PHASE[N−1] are prepared. However, schemesexist for reordering the phases in the stated order with respect to thefrequency domain. No limitation is intended in this regard. The N phasechanging values (or phase changing sets) may also change the phases ofblocks in the time domain or in the time-frequency domain to obtain asymbol arrangement as described in Embodiment 1. Although the aboveexamples discuss a phase changing scheme with a period (cycle) of N, thesame effects are obtainable using N phase changing values (or phasechanging sets) at random. That is, the N phase changing values (or phasechanging sets) need not always for a regular period (cycle). As long asthe above-described conditions are satisfied, great quality datareception improvements are realizable for the reception device.

Furthermore, given the existence of modes for spatial multiplexing MIMOschemes, MIMO schemes using a fixed precoding matrix, space-time blockcoding schemes, single-stream transmission, and schemes using a regularchange of phase (the transmission schemes described in Embodiments 1through 4), the transmission device (broadcaster, base station) mayselect any one of these transmission schemes.

As described in Non-Patent Literature 3, spatial multiplexing MIMOschemes involve transmitting signals s1 and s2, which are mapped using aselected modulation scheme, on each of two different antennas. Asdescribed in Embodiments 1 through 4, MIMO schemes using a fixedprecoding matrix involve performing precoding only (with no change ofphase). Further, space-time block coding schemes are described inNon-Patent Literature 9, 16, and 17. Single-stream transmission schemesinvolve transmitting signal s1, mapped with a selected modulationscheme, from an antenna after performing predetermined processing.

Schemes using multi-carrier transmission such as OFDM involve a firstcarrier group made up of a plurality of carriers and a second carriergroup made up of a plurality of carriers different from the firstcarrier group, and so on, such that multi-carrier transmission isrealized with a plurality of carrier groups. For each carrier group, anyof spatial multiplexing MIMO schemes, MIMO schemes using a fixedprecoding matrix, space-time block coding schemes, single-streamtransmission, and schemes using a regular change of phase may be used.In particular, schemes using a regular change of phase on a selected(sub-)carrier group are preferably used to realize the presentEmbodiment.

When a change of phase is performed, then for example, a phase changingvalue for PHASE[i] of X radians is performed on only one precodedbaseband signal, the phase changers of FIGS. 3, 4, 5, 12, 25, 29, 51,and 53 multiplies precoded baseband signal z2′ by e^(jX). Then, for achange of phase by, for example, a phase changing set for PHASE[i] of Xradians and Y radians is performed on both precoded baseband signals,the phase changers from FIGS. 26, 27, 28, 52, and 54 multiplies precodedbaseband signal z2′ by e^(jX) and multiplies precoded baseband signalz1′ by e^(jY).

Embodiment B1

The following describes a sample configuration of an application of thetransmission schemes and reception schemes discussed in the aboveembodiments and a system using the application.

FIG. 36 illustrates the configuration of a system that includes devicesexecuting transmission schemes and reception schemes described in theabove Embodiments. As shown in FIG. 36, the devices executingtransmission schemes and reception schemes described in the aboveEmbodiments include various receivers such as a broadcaster, atelevision 3611, a DVD recorder 3612, a STB (set-top box) 3613, acomputer 3620, a vehicle-mounted television 3641, a mobile phone 3630and so on within a digital broadcasting system 3600. Specifically, thebroadcaster 3601 uses a transmission scheme discussed in theabove-described Embodiments to transmit multiplexed data, in whichvideo, audio, and other data are multiplexed, over a predeterminedtransmission band.

The signals transmitted by the broadcaster 3601 are received by anantenna (such as antenna 3660 or 3640) embedded within or externallyconnected to each of the receivers. Each receiver obtains themultiplexed data by using reception schemes discussed in theabove-described Embodiments to demodulate the signals received by theantenna. Accordingly, the digital broadcasting system 3600 is able torealize the effects of the present invention, as discussed in theabove-described Embodiments.

The video data included in the multiplexed data are coded with a videocoding method compliant with a standard such as MPEG-2 (Moving PictureExperts Group), MPEG4-AVC (Advanced Video Coding), VC-1, or the like.The audio data included in the multiplexed data are encoded with anaudio coding method compliant with a standard such as Dolby AC-3 (AudioCoding), Dolby Digital Plus, MLP (Meridian Lossless Packing), DTS(Digital Theater Systems), DTS-HD, PCM (Pulse-Code Modulation), or thelike.

FIG. 37 illustrates the configuration of a receiver 7900 that executes areception scheme described in the above-described Embodiments. Thereceiver 3700 corresponds to a receiver included in one of thetelevision 3611, the DVD recorder 3612, the STB 3613, the computer 3620,the vehicle-mounted television 3641, the mobile phone 3630 and so onfrom FIG. 36. The receiver 3700 includes a tuner 3701 converting ahigh-frequency signal received by an antenna 3760 into a basebandsignal, and a demodulator 3702 demodulating the baseband signal soconverted to obtain the multiplexed data. The demodulator 3702 executesa reception scheme discussed in the above-described Embodiments, andthus achieves the effects of the present invention as explained above.

The receiver 3700 further includes a stream interface 3720 thatdemultiplexes the audio and video data in the multiplexed data obtainedby the demodulator 3702, a signal processor 3704 that decodes the videodata obtained from the demultiplexed video data into a video signal byapplying a video decoding method corresponding thereto and decodes theaudio data obtained from the demultiplexed audio data into an audiosignal by applying an audio decoding method corresponding thereto, anaudio output unit 3706 that outputs the decoded audio signal through aspeaker or the like, and a video display unit 3707 that outputs thedecoded video signal on a display or the like.

When, for example, a user uses a remote control 3750, information for aselected channel (selected (television) program or audio broadcast) istransmitted to an operation input unit 3710. Then, the receiver 3700performs processing on the received signal received by the antenna 3760that includes demodulating the signal corresponding to the selectedchannel, performing error-correcting decoding, and so on, in order toobtain the received data. At this point, the receiver 3700 obtainscontrol symbol information that includes information on the transmissionscheme (the transmission scheme, modulation scheme, error-correctionscheme, and so on from the above-described Embodiments) (as describedusing FIGS. 5 and 41) from control symbols included the signalcorresponding to the selected channel. As such, the receiver 3700 isable to correctly set the reception operations, demodulation scheme,error-correction scheme and so on, thus enabling the data included inthe data symbols transmitted by the broadcaster (base station) to beobtained. Although the above description is given for an example of theuser using the remote control 3750, the same operations apply when theuser presses a selection key embedded in the receiver 3700 to select achannel.

According to this configuration, the user is able to view programsreceived by the receiver 3700.

The receiver 3700 pertaining to the present Embodiment further includesa drive 3708 that may be a magnetic disk, an optical disc, anon-volatile semiconductor memory, or a similar recording medium. Thereceiver 3700 stores data included in the demultiplexed data obtainedthrough demodulation by the demodulator 3702 and error-correctingdecoding (in some circumstances, the data obtained through demodulationby the demodulator 3702 may not be subject to error correction. Also,the receiver 3700 may perform further processing after error correction.The same hereinafter applies to similar statements concerning othercomponents), data corresponding to such data (e.g., data obtainedthrough compression of such data), data obtained through audio and videoprocessing, and so on, on the drive 3708. Here, an optical disc is arecording medium, such as DVD (Digital Versatile Disc) or BD (Blu-rayDisc), that is readable and writable with the use of a laser beam. Amagnetic disk is a floppy disk, a hard disk, or similar recording mediumon which information is storable through the use of magnetic flux tomagnetize a magnetic body. A non-volatile semiconductor memory is arecording medium, such as flash memory or ferroelectric random accessmemory, composed of semiconductor element(s). Specific examples ofnon-volatile semiconductor memory include an SD card using flash memoryand a Flash SSD (Solid State Drive). Naturally, the specific types ofrecording media mentioned herein are merely examples. Other types ofrecording mediums may also be used.

According to this structure, the user is able to record and storeprograms received by the receiver 3700, and is thereby able to viewprograms at any given time after broadcasting by reading out therecorded data thereof.

Although the above explanations describe the receiver 3700 storingmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding on the drive 3708, a portion of the dataincluded in the multiplexed data may instead be extracted and recorded.For example, when data broadcasting services or similar content isincluded along with the audio and video data in the multiplexed dataobtained through demodulation by the demodulator 3702 anderror-correcting decoding, the audio and video data may be extractedfrom the multiplexed data demodulated by the demodulator 3702 and storedas new multiplexed data. Furthermore, the drive 3708 may store eitherthe audio data or the video data included in the multiplexed dataobtained through demodulation by the demodulator 3702 anderror-correcting decoding as new multiplexed data. The aforementioneddata broadcasting service content included in the multiplexed data mayalso be stored on the drive 3708.

Furthermore, when a television, recording device (e.g., a DVD recorder,BD recorder HDD recorder, SD card, or similar), or mobile phoneincorporating the receiver 3700 of the present invention receivesmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding that includes data for correcting bugs insoftware used to operate the television or recording device, forcorrecting bugs in software for preventing personal information andrecorded data from being leaked, and so on, such software bugs may becorrected by installing the data on the television or recording device.As such, bugs in the receiver 3700 are corrected through the inclusionof data for correcting bugs in the software of the receiver 3700.Accordingly, the television, recording device, or mobile phoneincorporating the receiver 3700 may be made to operate more reliably.

Here, the process of extracting a portion of the data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding is performed by, for example, the streaminterface 3703. Specifically, the stream interface 3703, demultiplexesthe various data included in the multiplexed data demodulated by thedemodulator 3702, such as audio data, video data, data broadcastingservice content, and so on, as instructed by a non-diagrammed controllersuch as a CPU. The stream interface 3703 then extracts and multiplexesonly the indicated demultiplexed data, thus generating new multiplexeddata. The data to be extracted from the demultiplexed data may bedetermined by the user or may be determined in advance according to thetype of recording medium.

According to such a structure, the receiver 3700 is able to extract andrecord only the data needed in order to view the recorded program. Assuch, the amount of data to be recorded can be reduced.

Although the above explanation describes the drive 3708 as storingmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding, the video data included in themultiplexed data so obtained may be converted by using a different videocoding method than the original video coding method applied thereto, soas to reduce the amount of data or the bit rate thereof. The drive 3708may then store the converted video data as new multiplexed data. Here,the video coding method used to generate the new video data may conformto a different standard than that used to generate the original videodata. Alternatively, the same video coding method may be used withdifferent parameters. Similarly, the audio data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding may be converted by using a differentaudio coding method than the original audio coding method appliedthereto, so as to reduce the amount of data or the bit rate thereof. Thedrive 3708 may then store the converted audio data as new multiplexeddata.

Here, the process by which the audio or video data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding is converted so as to reduce the amount ofdata or the bit rate thereof is performed by, for example, the streaminterface 3703 or the signal processor 3704. Specifically, the streaminterface 3703 demultiplexes the various data included in themultiplexed data demodulated by the demodulator 3702, such as audiodata, video data, data broadcasting service content, and so on, asinstructed by an undiagrammed controller such as a CPU. The signalprocessor 3704 then performs processing to convert the video data sodemultiplexed by using a different video coding method than the originalvideo coding method applied thereto, and performs processing to convertthe audio data so demultiplexed by using a different video coding methodthan the original audio coding method applied thereto. As instructed bythe controller, the stream interface 3703 then multiplexes the convertedaudio and video data, thus generating new multiplexed data. The signalprocessor 3704 may, in accordance with instructions from the controller,performing conversion processing on either the video data or the audiodata, alone, or may perform conversion processing on both types of data.In addition, the amounts of video data and audio data or the bit ratethereof to be obtained by conversion may be specified by the user ordetermined in advance according to the type of recording medium.

According to such a structure, the receiver 3700 is able to modify theamount of data or the bitrate of the audio and video data for storageaccording to the data storage capacity of the recording medium, oraccording to the data reading or writing speed of the drive 3708.Therefore, programs can be stored on the drive despite the storagecapacity of the recording medium being less than the amount ofmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding, or the data reading or writing speed ofthe drive being lower than the bit rate of the demultiplexed dataobtained through demodulation by the demodulator 3702. As such, the useris able to view programs at any given time after broadcasting by readingout the recorded data.

The receiver 3700 further includes a stream output interface 3709 thattransmits the multiplexed data demultiplexed by the demodulator 3702 toexternal devices through a communications medium 3730. The stream outputinterface 3709 may be, for example, a wireless communication devicetransmitting modulated multiplexed data to an external device using awireless transmission scheme conforming to a wireless communicationstandard such as Wi-Fi™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.11g, IEEE802.11n, and so on), WiGig, WirelessHD, Bluetooth, ZigBee, and so onthrough a wireless medium (corresponding to the communications medium3730). The stream output interface 3709 may also be a wiredcommunication device transmitting modulated multiplexed data to anexternal device using a communication scheme conforming to a wiredcommunication standard such as Ethernet™, USB (Universal Serial Bus),PLC (Power Line Communication), HDMI (High-Definition MultimediaInterface) and so on through a wired transmission path (corresponding tothe communications medium 3730) connected to the stream output interface3709.

According to this configuration, the user is able to use an externaldevice with the multiplexed data received by the receiver 3700 using thereception scheme described in the above-described Embodiments. The usageof multiplexed data by the user here includes use of the multiplexeddata for real-time viewing on an external device, recording of themultiplexed data by a recording unit included in an external device, andtransmission of the multiplexed data from an external device to a yetanother external device.

Although the above explanations describe the receiver 3700 outputtingmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding through the stream output interface 3709,a portion of the data included in the multiplexed data may instead beextracted and output. For example, when data broadcasting services orsimilar content is included along with the audio and video data in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding, the audio and video data may be extractedfrom the multiplexed data obtained through demodulation by thedemodulator 3702 and error-correcting decoding, multiplexed and outputby the stream output interface 3709 as new multiplexed data. Inaddition, the stream output interface 3709 may store either the audiodata or the video data included in the multiplexed data obtained throughdemodulation by the demodulator 3702 and error-correcting decoding asnew multiplexed data.

Here, the process of extracting a portion of the data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding is performed by, for example, the streaminterface 3703. Specifically, the stream interface 3703 demultiplexesthe various data included in the multiplexed data demodulated by thedemodulator 3702, such as audio data, video data, data broadcastingservice content, and so on, as instructed by an undiagrammed controllersuch as a CPU. The stream interface 3703 then extracts and multiplexesonly the indicated demultiplexed data, thus generating new multiplexeddata. The data to be extracted from the demultiplexed data may bedetermined by the user or may be determined in advance according to thetype of stream output interface 3709.

According to this structure, the receiver 3700 is able to extract andoutput only the required data to an external device. As such, fewermultiplexed data are output using less communication bandwidth.

Although the above explanation describes the stream output interface3709 as outputting multiplexed data obtained through demodulation by thedemodulator 3702 and error-correcting decoding, the video data includedin the multiplexed data so obtained may be converted by using adifferent video coding method than the original video coding methodapplied thereto, so as to reduce the amount of data or the bit ratethereof. The stream output interface 3709 may then output the convertedvideo data as new multiplexed data. Here, the video coding method usedto generate the new video data may conform to a different standard thanthat used to generate the original video data. Alternatively, the samevideo coding method may be used with different parameters. Similarly,the audio data included in the multiplexed data obtained throughdemodulation by the demodulator 3702 and error-correcting decoding maybe converted by using a different audio coding method than the originalaudio coding method applied thereto, so as to reduce the amount of dataor the bit rate thereof. The stream output interface 3709 may thenoutput the converted audio data as new multiplexed data.

Here, the process by which the audio or video data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding is converted so as to reduce the amount ofdata or the bit rate thereof is performed by, for example, the streaminterface 3703 or the signal processor 3704. Specifically, the streaminterface 3703 demultiplexes the various data included in themultiplexed data demodulated by the demodulator 3702, such as audiodata, video data, data broadcasting service content, and so on, asinstructed by an undiagrammed controller. The signal processor 3704 thenperforms processing to convert the video data so demultiplexed by usinga different video coding method than the original video coding methodapplied thereto, and performs processing to convert the audio data sodemultiplexed by using a different video coding method than the originalaudio coding method applied thereto. As instructed by the controller,the stream interface 3703 then multiplexes the converted audio and videodata, thus generating new multiplexed data. The signal processor 3704may, in accordance with instructions from the controller, performingconversion processing on either the video data or the audio data, alone,or may perform conversion processing on both types of data. In addition,the amounts of video data and audio data or the bit rate thereof to beobtained by conversion may be specified by the user or determined inadvance according to the type of stream output interface 3709.

According to this structure, the receiver 3700 is able to modify the bitrate of the video and audio data for output according to the speed ofcommunication with the external device. Thus, despite the speed ofcommunication with an external device being slower than the bit rate ofthe multiplexed data obtained through demodulation by the demodulator3702 and error-correcting decoding, by outputting new multiplexed datafrom the stream output interface to the external device, the user isable to use the new multiplexed data with other communication devices.

The receiver 3700 further includes an audiovisual output interface 3711that outputs audio and video signals decoded by the signal processor3704 to the external device through an external communications medium.The audiovisual output interface 3711 may be, for example, a wirelesscommunication device transmitting modulated audiovisual data to anexternal device using a wireless transmission scheme conforming to awireless communication standard such as Wi-Fi™ (IEEE 802.11a, IEEE802.11b, IEEE 802.11g, IEEE 802.11n, and so on), WiGig, WirelessHD,Bluetooth, ZigBee, and so on through a wireless medium. The streamoutput interface 3709 may also be a wired communication devicetransmitting modulated audiovisual data to an external device using acommunication scheme conforming to a wired communication standard suchas Ethernet™, USB, PLC, HDMI, and so on through a wired transmissionpath connected to the stream output interface 3709. Furthermore, thestream output interface 3709 may be a terminal for connecting a cablethat outputs analogue audio signals and video signals as-is.

According to such a structure, the user is able to use the audio signalsand video signals decoded by the signal processor 3704 with an externaldevice.

Further, the receiver 3700 includes an operation input unit 3710 thatreceives user operations as input. The receiver 3700 behaves inaccordance with control signals input by the operation input unit 3710according to user operations, such as by switching the power supply ONor OFF, changing the channel being received, switching subtitle displayON or OFF, switching between languages, changing the volume output bythe audio output unit 3706, and various other operations, includingmodifying the settings for receivable channels and the like.

The receiver 3700 may further include functionality for displaying anantenna level representing the received signal quality while thereceiver 3700 is receiving a signal. The antenna level may be, forexample, a index displaying the received signal quality calculatedaccording to the RSSI (Received Signal Strength Indicator), the receivedsignal magnetic field strength, the C/N (carrier-to-noise) ratio, theBER, the packet error rate, the frame error rate, the channel stateinformation, and so on, received by the receiver 3700 and indicating thelevel and the quality of a received signal. In such circumstances, thedemodulator 3702 includes a signal quality calibrator that measures theRSSI, the received signal magnetic field strength, the C/N ratio, theBER, the packet error rate, the frame error rate, the channel stateinformation, and so on. In response to user operations, the receiver3700 displays the antenna level (signal level, signal quality) in auser-recognizable format on the video display unit 3707. The displayformat for the antenna level (signal level, signal quality) may be anumerical value displayed according to the RSSI, the received signalmagnetic field strength, the C/N ratio, the BER, the packet error rate,the frame error rate, the channel state information, and so on, or maybe an image display that varies according to the RSSI, the receivedsignal magnetic field strength, the C/N ratio, the BER, the packet errorrate, the frame error rate, the channel state information, and so on.The receiver 3700 may display multiple antenna level (signal level,signal quality) calculated for each stream s1, s2, and so ondemultiplexed using the reception scheme discussed in theabove-described Embodiments, or may display a single antenna level(signal level, signal quality) calculated for all such streams. When thevideo data and audio data composing a program are transmittedhierarchically, the signal level (signal quality) may also be displayedfor each hierarchical level.

According to the above structure, the user is given an understanding ofthe antenna level (signal level, signal quality) numerically or visuallyduring reception using the reception schemes discussed in theabove-described Embodiments.

Although the above example describes the receiver 3700 as including theaudio output unit 3706, the video display unit 3707, the drive 3708, thestream output interface 3709, and the audiovisual output interface 3711,all of these components are not strictly necessary. As long as thereceiver 3700 includes at least one of the above-described components,the user is able to use the multiplexed data obtained throughdemodulation by the demodulator 3702 and error-correcting decoding. Anyreceiver may be freely combined with the above-described componentsaccording to the usage scheme.

(Multiplexed Data)

The following is a detailed description of a sample configuration ofmultiplexed data. The data configuration typically used in broadcastingis an MPEG-2 transport stream (TS). Therefore the following descriptiondescribes an example related to MPEG2-TS. However, the dataconfiguration of the multiplexed data transmitted by the transmissionand reception schemes discussed in the above-described Embodiments isnot limited to MPEG2-TS. The advantageous effects of the above-describedEmbodiments are also achievable using any other data structure.

FIG. 38 illustrates a sample configuration for multiplexed data. Asshown, the multiplexed data are elements making up programmes (orevents, being a portion thereof) currently provided by various services.For example, one or more video streams, audio streams, presentationgraphics (PG) streams, interactive graphics (IG) streams, and other suchelement streams are multiplexed to obtain the multiplexed data. When abroadcast program provided by the multiplexed data is a movie, the videostreams represent main video and sub video of the movie, the audiostreams represent main audio of the movie and sub-audio to be mixed withthe main audio, and the presentation graphics streams representsubtitles for the movie. Main video refers to video images normallypresented on a screen, whereas sub-video refers to video images (forexample, images of text explaining the outline of the movie) to bepresented in a small window inserted within the video images. Theinteractive graphics streams represent an interactive display made up ofGUI (Graphical User Interface) components presented on a screen.

Each stream included in the multiplexed data is identified by anidentifier, termed a PID, uniquely assigned to the stream. For example,PID 0x1011 is assigned to the video stream used for the main video ofthe movie, PIDs 0x1100 through 0x111F are assigned to the audio streams,PIDs 0x1200 through 0x121F are assigned to the presentation graphics,PIDs 0x1400 through 0x141F are assigned to the interactive graphics,PIDs 0x1B00 through 0x1B1F are assigned to the video streams used forthe sub-video of the movie, and PIDs 0x1A00 through 0x1A1F are assignedto the audio streams used as sub-audio to be mixed with the main audioof the movie.

FIG. 39 is a schematic diagram illustrating an example of themultiplexed data being multiplexed. First, a video stream 3901, made upof a plurality of frames, and an audio stream 3904, made up of aplurality of audio frames, are respectively converted into PES packetsequence 3902 and 3905, then further converted into TS packets 3903 and3906. Similarly, a presentation graphics stream 3911 and an interactivegraphics stream 3914 are respectively converted into PES packet sequence3912 and 3915, then further converted into TS packets 3913 and 3916. Themultiplexed data 3917 is made up of the TS packets 3903, 3906, 3913, and3916 multiplexed into a single stream.

FIG. 40 illustrates further details of a PES packet sequence ascontained in the video stream. The first tier of FIG. 40 shows a videoframe sequence in the video stream. The second tier shows a PES packetsequence. Arrows yy1, yy2, yy3, and yy4 indicate the plurality of VideoPresentation Units, which are I-pictures, B-pictures, and P-pictures, inthe video stream as divided and individually stored as the payload of aPES packet. Each PES packet has a PES header. A PES header contains aPTS (Presentation Time Stamp) at which the picture is to be displayed, aDTS (Decoding Time Stamp) at which the picture is to be decoded, and soon.

FIG. 41 illustrates the structure of a TS packet as ultimately writteninto the multiplexed data. A TS packet is a 188-byte fixed-length packetmade up of a 4-byte PID identifying the stream and of a 184-byte TSpayload containing the data. The above-described PES packets are dividedand individually stored as the TS payload. For a BD-ROM, each TS packethas a 4-byte TP_Extra_Header affixed thereto to build a 192-byte sourcepacket, which is to be written as the multiplexed data. TheTP_Extra_Header contains information such as an Arrival_Time_Stamp(ATS). The ATS indicates a time for starring transfer of the TS packetto the PID filter of a decoder. The multiplexed data are made up ofsource packets arranged as indicated in the bottom tier of FIG. 41. ASPN (source packet number) is incremented for each packet, beginning atthe head of the multiplexed data.

In addition to the video streams, audio streams, presentation graphicsstreams, and the like, the TS packets included in the multiplexed dataalso include a PAT (Program Association Table), a PMT (Program MapTable), a PCR (Program Clock Reference) and so on. The PAT indicates thePID of a PMT used in the multiplexed data, and the PID of the PAT itselfis registered as 0. The PMT includes PIDs identifying the respectivestreams, such as video, audio and subtitles, contained in themultiplexed data and attribute information (frame rate, aspect ratio,and the like) of the streams identified by the respective PIDs. Inaddition, the PMT includes various types of descriptors relating to themultiplexed data. One such descriptor may be copy control informationindicating whether or not copying of the multiplexed data is permitted.The PCR includes information for synchronizing the ATC (Arrival TimeClock) serving as the chronological axis of the ATS to the STC (SystemTime Clock) serving as the chronological axis of the PTS and DTS. EachPCR packet includes an STC time corresponding to the ATS at which thepacket is to be transferred to the decoder.

FIG. 42 illustrates the detailed data configuration of a PMT. The PMTstarts with a PMT header indicating the length of the data contained inthe PMT. Following the PMT header, descriptors pertaining to themultiplexed data are arranged. One example of a descriptor included inthe PMT is the copy control information described above. Following thedescriptors, stream information pertaining to the respective streamsincluded in the multiplexed data is arranged. Each piece of streaminformation is composed of stream descriptors indicating a stream typeidentifying a compression codec employed for a corresponding stream, aPID for the stream, and attribute information (frame rate, aspect ratio,and the like) of the stream. The PMT includes the same number of streamdescriptors as the number of streams included in the multiplexed data.

When recorded onto a recoding medium or the like, the multiplexed dataare recorded along with a multiplexed data information file.

FIG. 43 illustrates a sample configuration for the multiplexed datainformation file. As shown, the multiplexed data information file ismanagement information for the multiplexed data, is provided inone-to-one correspondence with the multiplexed data, and is made up ofmultiplexed data information, stream attribute information, and an entrymap.

The multiplexed data information is made up of a system rate, a playbackstart time, and a playback end time. The system rate indicates themaximum transfer rate of the multiplexed data to the PID filter of alater-described system target decoder. The multiplexed data includes ATSat an interval set so as not to exceed the system rate. The playbackstart time is set to the time specified by the PTS of the first videoframe in the multiplexed data, whereas the playback end time is set tothe time calculated by adding the playback duration of one frame to thePTS of the last video frame in the multiplexed data.

FIG. 44 illustrates a sample configuration for the stream attributeinformation included in the multiplexed data information file. As shown,the stream attribute information is attribute information for eachstream included in the multiplexed data, registered for each PID. Thatis, different pieces of attribute information are provided for differentstreams, namely for the video streams, the audio streams, thepresentation graphics streams, and the interactive graphics streams. Thevideo stream attribute information indicates the compression codecemployed to compress the video stream, the resolution of individualpictures constituting the video stream, the aspect ratio, the framerate, and so on. The audio stream attribute information indicates thecompression codec employed to compress the audio stream, the number ofchannels included in the audio stream, the language of the audio stream,the sampling frequency, and so on. This information is used toinitialize the decoder before playback by a player.

In the present Embodiment, the stream type included in the PMT is usedamong the information included in the multiplexed data. When themultiplexed data are recorded on a recording medium, the video streamattribute information included in the multiplexed data information fileis used. Specifically, the video coding method and device described inany of the above Embodiments may be modified to additionally include astep or unit of setting a specific piece of information in the streamtype included in the PMT or in the video stream attribute information.The specific piece of information is for indicating that the video dataare generated by the video coding method and device described in theEmbodiment. According to such a structure, video data generated by thevideo coding method and device described in any of the above Embodimentsis distinguishable from video data compliant with other standards.

FIG. 45 illustrates a sample configuration of an audiovisual outputdevice 4500 that includes a reception device 4504 receiving a modulatedsignal that includes audio and video data transmitted by a broadcaster(base station) or data intended for broadcasting. The configuration ofthe reception device 4504 corresponds to the reception device 3700 fromFIG. 37. The audiovisual output device 4500 incorporates, for example,an OS (Operating System), or incorporates a communication device 4506for connecting to the Internet (e.g., a communication device intendedfor a wireless LAN (Local Area Network) or for Ethernet™). As such, avideo display unit 4501 is able to simultaneously display audio andvideo data, or video in video data for broadcast 4502, and hypertext4503 (from the World Wide Web) provided over the Internet. By operatinga remote control 4507 (alternatively, a mobile phone or keyboard),either of the video in video data for broadcast 4502 and the hypertext4503 provided over the Internet may be selected to change operations.For example, when the hypertext 4503 provided over the Internet isselected, the website displayed may be changed by remote controloperations. When audio and video data, or video in video data forbroadcast 4502 is selected, information from a selected channel(selected (television) program or audio broadcast) may be transmitted bythe remote control 4507. As such, an interface 4505 obtains theinformation transmitted by the remote control. The reception device 4504performs processing such as demodulation and error-correctioncorresponding to the selected channel, thereby obtaining the receiveddata. At this point, the reception device 4504 obtains control symbolinformation that includes information on the transmission scheme (asdescribed using FIG. 5) from control symbols included the signalcorresponding to the selected channel. As such, the reception device4504 is able to correctly set the reception operations, demodulationscheme, error-correction scheme and so on, thus enabling the dataincluded in the data symbols transmitted by the broadcaster (basestation) to be obtained. Although the above description is given for anexample of the user using the remote control 4507, the same operationsapply when the user presses a selection key embedded in the audiovisualoutput device 4500 to select a channel.

In addition, the audiovisual output device 4500 may be operated usingthe Internet. For example, the audiovisual output device 4500 may bemade to record (store) a program through another terminal connected tothe Internet. (Accordingly, the audiovisual output device 4500 shouldinclude the drive 3708 from FIG. 37.) The channel is selected beforerecording begins. As such, the reception device 4504 performs processingsuch as demodulation and error-correction corresponding to the selectedchannel, thereby obtaining the received data. At this point, thereception device 4504 obtains control symbol information that includesinformation on the transmission scheme (the transmission scheme,modulation scheme, error-correction scheme, and so on from theabove-described Embodiments) (as described using FIG. 5) from controlsymbols included the signal corresponding to the selected channel. Assuch, the reception device 4504 is able to correctly set the receptionoperations, demodulation scheme, error-correction scheme and so on, thusenabling the data included in the data symbols transmitted by thebroadcaster (base station) to be obtained.

(Supplement)

The present description considers a communications/broadcasting devicesuch as a broadcaster, a base station, an access point, a terminal, amobile phone, or the like provided with the transmission device, and acommunications device such as a television, radio, terminal, personalcomputer, mobile phone, access point, base station, or the like providedwith the reception device. The transmission device and the receptiondevice pertaining to the present invention are communication devices ina form able to execute applications, such as a television, radio,personal computer, mobile phone, or similar, through connection to somesort of interface (e.g., USB).

Furthermore, in the present Embodiment, symbols other than data symbols,such as pilot symbols (namely preamble, unique word, postamble,reference symbols, scattered pilot symbols and so on), symbols intendedfor control information, and so on may be freely arranged within theframe. Although pilot symbols and symbols intended for controlinformation are presently named, such symbols may be freely namedotherwise as the function thereof remains the important consideration.

Provided that a pilot symbol, for example, is a known symbol modulatedwith PSK modulation in the transmitter and receiver (alternatively, thereceiver may be synchronized such that the receiver knows the symbolstransmitted by the transmitter), the receiver is able to use this symbolfor frequency synchronization, time synchronization, channel estimation(CSI (Channel State Information) estimation for each modulated signal),signal detection, and the like.

The symbols intended for control information are symbols transmittinginformation (such as the modulation scheme, error-correcting codingscheme, coding rate of error-correcting codes, and setting informationfor the top layer used in communications) transmitted to the receivingparty in order to execute transmission of non-data (i.e., applications).

The present invention is not limited to the Embodiments, but may also berealized in various other ways. For example, while the above Embodimentsdescribe communication devices, the present invention is not limited tosuch devices and may be implemented as software for the correspondingcommunications scheme.

Although the above-described Embodiments describe phase changing schemesfor schemes of transmitting two modulated signals from two antennas, nolimitation is intended in this regard. Precoding and a change of phasemay be performed on four signals that have been mapped to generate fourmodulated signals transmitted using four antennas. That is, the presentinvention is applicable to performing a change of phase on N signalsthat have been mapped and precoded to generate N modulated signalstransmitted using N antennas.

Although the above-described Embodiments describe examples of systemswhere two modulated signals are transmitted from two antennas andreceived by two respective antennas in a MIMO system, the presentinvention is not limited in this regard and is also applicable to MISO(Multiple Input Single Output) systems. In a MISO system, the receptiondevice does not include antenna 701_Y, wireless unit 703_Y, channelfluctuation estimator 707_1 for modulated signal z1, and channelfluctuation estimator 707_2 for modulated signal z2 from FIG. 7.However, the processing described in Embodiment 1 may still be executedto estimate r1 and r2. Technology for receiving and decoding a pluralityof signals transmitted simultaneously at a common frequency are receivedby a single antenna is widely known. The present invention is additionalprocessing supplementing conventional technology for a signal processorreverting a phase changed by the transmitter.

Although the present invention describes examples of systems where twomodulated signals are transmitted from two antennas and received by tworespective antennas in a MIMO system, the present invention is notlimited in this regard and is also applicable to MISO systems. In a MISOsystem, the transmission device performs precoding and change of phasesuch that the points described thus far are applicable. However, thereception device does not include antenna 701_Y, wireless unit 703_Y,channel fluctuation estimator 707_1 for modulated signal z1, and channelfluctuation estimator 707_2 for modulated signal z2 from FIG. 7.However, the processing described in the present description may stillbe executed to estimate the data transmitted by the transmission device.Technology for receiving and decoding a plurality of signals transmittedsimultaneously at a common frequency are received by a single antenna iswidely known (a single-antenna receiver may apply ML operations (Max-logAPP or similar)). The present invention may have the signal processor711 from FIG. 7 perform demodulation (detection) by taking the precodingand change of phase applied by the transmitter into consideration.

The present description uses terms such as precoding, precoding weights,precoding matrix, and so on. The terminology itself may be otherwise(e.g., may be alternatively termed a codebook) as the key point of thepresent invention is the signal processing itself.

Furthermore, although the present description discusses examples mainlyusing OFDM as the transmission scheme, the invention is not limited inthis manner. Multi-carrier schemes other than OFDM and single-carrierschemes may all be used to achieve similar Embodiments. Here,spread-spectrum communications may also be used. When single-carrierschemes are used, a change of phase is performed with respect to thetime domain.

In addition, although the present description discusses the use of MLoperations, APP, Max-log APP, ZF, MMSE and so on by the receptiondevice, these operations may all be generalized as wave detection,demodulation, detection, estimation, and demultiplexing as the softresults (log-likelihood and log-likelihood ratio) and the hard results(zeroes and ones) obtained thereby are the individual bits of datatransmitted by the transmission device.

Different data may be transmitted by each stream s1(t) and s2(t) (s1(i),s2(i)), or identical data may be transmitted thereby.

The two stream baseband signals s1(i) and s2(i) (where i indicatessequence (with respect to time or (carrier) frequency)) undergoprecoding and a regular change of phase (the order of operations may befreely reversed) to generate two post-processing baseband signals z1(i)and z2(i). For post-processing baseband signal z1(i), the in-phasecomponent I is I₁(i) while the quadrature component is Q₁(i), and forpost processing baseband signal z2(i), the in-phase component is I₁(i)while the quadrature component is Q₂(i). The baseband components may beswitched, as long as the following holds.

Let the in-phase component and the quadrature component of switchedbaseband signal r1(i) be I₁(i) and Q₂(i), and the in-phase component andthe quadrature component of switched baseband signal r2(i) be I₂(i) andQ₁(i). The modulated signal corresponding to switched baseband signalr1(i) is transmitted by transmit antenna 1 and the modulated signalcorresponding to switched baseband signal r2(i) is transmitted fromtransmit antenna 2, simultaneously on a common frequency. As such, themodulated signal corresponding to switched baseband signal r1(i) and themodulated signal corresponding to switched baseband signal r2(i) aretransmitted from different antennas, simultaneously on a commonfrequency. Alternatively,

For switched baseband signal r1(i), the in-phase component may be I₁(i)while the quadrature component may be I₂(i), and for switched basebandsignal r2(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be Q₂(i).

For switched baseband signal r1(i), the in-phase component may be I₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r2(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be Q₂(i).

For switched baseband signal r1(i), the in-phase component may be I₁(i)while the quadrature component may be I₂(i), and for switched basebandsignal r2(i), the in-phase component may be Q₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r1(i), the in-phase component may be I₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r2(i), the in-phase component may be Q₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r1(i), the in-phase component may be I₁(i)while the quadrature component may be Q₂(i), and for switched basebandsignal r2(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be I₂(i).

For switched baseband signal r1(i), the in-phase component may be Q₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r2(i), the in-phase component may be I₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r1(i), the in-phase component may be Q₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r2(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be I₂(i).

For switched baseband signal r2(i), the in-phase component may be I₁(i)while the quadrature component may be I₂(i), and for switched basebandsignal r1(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be Q₂(i).

For switched baseband signal r2(i), the in-phase component may be I₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r1(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be Q₂(i).

For switched baseband signal r2(i), the in-phase component may be I₁(i)while the quadrature component may be I₂(i), and for switched basebandsignal r1(i), the in-phase component may be Q₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r2(i), the in-phase component may be I₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r1(i), the in-phase component may be Q₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r2(i), the in-phase component may be I₁(i)while the quadrature component may be Q₂(i), and for switched basebandsignal r1(i), the in-phase component may be I₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r2(i), the in-phase component may be I₁(i)while the quadrature component may be Q₂(i), and for switched basebandsignal r1(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be I₂(i).

For switched baseband signal r2(i), the in-phase component may be Q₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r1(i), the in-phase component may be I₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r2(i), the in-phase component may be Q₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r1(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be I₂(i). Alternatively, although the above descriptiondiscusses performing two types of signal processing on both streamsignals so as to switch the in-phase component and quadrature componentof the two signals, the invention is not limited in this manner. The twotypes of signal processing may be performed on more than two streams, soas to switch the in-phase component and quadrature component thereof.

Alternatively, although the above examples describe switching basebandsignals having a common time (common (sub-)carrier) frequency), thebaseband signals being switched need not necessarily have a common time.For example, any of the following are possible.

For switched baseband signal r1(i), the in-phase component may beI₁(i+v) while the quadrature component may be Q₂(i+w), and for switchedbaseband signal r2(i), the in-phase component may be I₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r1(i), the in-phase component may beI₁(i+v) while the quadrature component may be I₂(i+w), and for switchedbaseband signal r2(i), the in-phase component may be Q₁(i+v) while thequadrature component may be Q₂(i+w).

For switched baseband signal r1(i), the in-phase component may beI₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r2(i), the in-phase component may be Q₁(i+v) while thequadrature component may be Q₂(i+w).

For switched baseband signal r1(i), the in-phase component may beI₁(i+v) while the quadrature component may be I₂(i+w), and for switchedbaseband signal r2(i), the in-phase component may be Q₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r1(i), the in-phase component may beI₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r2(i), the in-phase component may be Q₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r1(i), the in-phase component may beI₁(i+v) while the quadrature component may be Q₂(i+w), and for switchedbaseband signal r2(i), the in-phase component may be Q₁(i+v) while thequadrature component may be I₂(i+w).

For switched baseband signal r1(i), the in-phase component may beQ₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r2(i), the in-phase component may be I₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r1(i), the in-phase component may beQ₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r2(i), the in-phase component may be Q₁(i+v) while thequadrature component may be I₂(i+w).

For switched baseband signal r2(i), the in-phase component may beI₁(i+v) while the quadrature component may be I₂(i+w), and for switchedbaseband signal r1(i), the in-phase component may be Q₁(i+v) while thequadrature component may be Q₂(i+w).

For switched baseband signal r2(i), the in-phase component may beI₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r1(i), the in-phase component may be Q₁(i+v) while thequadrature component may be Q₂(i+w).

For switched baseband signal r2(i), the in-phase component may beI₁(i+v) while the quadrature component may be I₂(i+w), and for switchedbaseband signal r1(i), the in-phase component may be Q₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r2(i), the in-phase component may beI₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r1(i), the in-phase component may be Q₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r2(i), the in-phase component may beI₁(i+v) while the quadrature component may be Q₂(i+w), and for switchedbaseband signal r1(i), the in-phase component may be I₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r2(i), the in-phase component may beI₁(i+v) while the quadrature component may be Q₂(i+w), and for switchedbaseband signal r1(i), the in-phase component may be Q₁(i+v) while thequadrature component may be I₂(i+w).

For switched baseband signal r2(i), the in-phase component may beQ₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r1(i), the in-phase component may be I₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r2(i), the in-phase component may beQ₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r1(i), the in-phase component may be Q₁(i+v) while thequadrature component may be I₂(i+w).

FIG. 55 illustrates a baseband signal switcher 5502 explaining theabove. As shown, of the two processed baseband signals z1(i) 5501_1 andz2(i) 5501_2, processed baseband signal z1(i) 5501_1 has in-phasecomponent I₁(i) and quadrature component Q₁(i), while processed basebandsignal z2(i) 5501_2 has in-phase component I₂(i) and quadraturecomponent Q₂(i). Then, after switching, switched baseband signal r1 (i)5503_1 has in-phase component I_(r1)(i) and quadrature componentQ_(r1)(i), while switched baseband signal r2(i) 5503_2 has in-phasecomponent I_(r2)(i) and quadrature component Q_(r2)(i). The in-phasecomponent I_(r1)(i) and quadrature component Q_(r1)(i) of switchedbaseband signal r1(i) 5503_1 and the in-phase component Ir2(i) andquadrature component Q_(r2)(i) of switched baseband signal r2(i) 5503_2may be expressed as any of the above. Although this example describesswitching performed on baseband signals having a common time (common((sub-)carrier) frequency) and having undergone two types of signalprocessing, the same may be applied to baseband signals having undergonetwo types of signal processing but having different times (different((sub-)carrier) frequencies).

Each of the transmit antennas of the transmission device and each of thereceive antennas of the reception device shown in the figures may beformed by a plurality of antennas.

The present description uses the symbol ∀, which is the universalquantifier, and the symbol ∃, which is the existential quantifier.

Furthermore, the present description uses the radian as the unit ofphase in the complex plane, e.g., for the argument thereof.

When dealing with the complex plane, the coordinates of complex numbersare expressible by way of polar coordinates. For a complex number z=a+jb(where a and b are real numbers and j is the imaginary unit), thecorresponding point (a, b) on the complex plane is expressed with thepolar coordinates [r, θ], converted as follows:

a=r×cos θb=r×sin θ

[Math. 49]

r=√{square root over (a ² +b ²)}  (formula 49)

where r is the absolute value of z (r=|z|), and θ is the argumentthereof. As such, z=a+jb is expressible as re^(jθ).

In the present invention, the baseband signals s1, s2, z1, and z2 aredescribed as being complex signals. A complex signal made up of in-phasesignal I and quadrature signal Q is also expressible as complex signalI+jQ. Here, either of I and Q may be equal to zero.

FIG. 46 illustrates a sample broadcasting system using the phasechanging scheme described in the present description. As shown, a videoencoder 4601 takes video as input, performs video encoding, and outputsencoded video data 4602. An audio encoder takes audio as input, performsaudio encoding, and outputs encoded audio data 4604. A data encoder 4605takes data as input, performs data encoding (e.g., data compression),and outputs encoded data 4606. Taken as a whole, these components form asource information encoder 4600.

A transmitter 4607 takes the encoded video data 4602, the encoded audiodata 4604, and the encoded data 4606 as input, performs error-correctingcoding, modulation, precoding, and phase changing (e.g., the signalprocessing by the transmission device from FIG. 3) on a subset of or onthe entirety of these, and outputs transmit signals 4608_1 through4608_N. Transmit signals 4608_1 through 4608_N are then transmitted byantennas 4609_1 through 4609_N as radio waves.

A receiver 4612 takes received signals 4611_1 through 4611_M received byantennas 4610_1 through 4610_M as input, performs processing such asfrequency conversion, change of phase, decoding of the precoding,log-likelihood ratio calculation, and error-correcting decoding (e.g.,the processing by the reception device from FIG. 7), and outputsreceived data 4613, 4615, and 4617. A source information decoder 4619takes the received data 4613, 4615, and 4617 as input. A video decoder4614 takes received data 4613 as input, performs video decoding, andoutputs a video signal. The video is then displayed on a televisiondisplay. An audio decoder 4616 takes received data 4615 as input. Theaudio decoder 4616 performs audio decoding and outputs an audio signal.the audio is then played through speakers. A data decoder 4618 takesreceived data 4617 as input, performs data decoding, and outputsinformation.

In the above-described Embodiments pertaining to the present invention,the number of encoders in the transmission device using a multi-carriertransmission scheme such as OFDM may be any number, as described above.Therefore, as in FIG. 4, for example, the transmission device may haveonly one encoder and apply a scheme for distributing output to themulti-carrier transmission scheme such as OFDM. In such circumstances,the wireless units 310A and 310B from FIG. 4 should replace theOFDM-related processors 1301A and 1301B from FIG. 12. The description ofthe OFDM-related processors is as given for Embodiment 1.

Although Embodiment 1 gives Math. 36 (formula 36) as an example of aprecoding matrix, another precoding matrix may also be used, when thefollowing scheme is applied.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 50} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}e^{j\; 0} & {\alpha \times e^{j\; \pi}} \\{\alpha \times e^{j\; 0}} & e^{j\; 0}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 50} \right)\end{matrix}$

In the precoding matrices of Math. 36 (formula 36) and Math. 50 (formula50), the value of a is set as given by Math. 37 (formula 37) and Math.38 (formula 38). However, no limitation is intended in this manner. Asimple precoding matrix is obtainable by setting α=1, which is also avalid value.

In Embodiment A1, the phase changers from FIGS. 3, 4, 6, 12, 25, 29, 51,and 53 are indicated as having a phase changing value of PHASE[i] (wherei=0, 1, 2 . . . N−2, N−1 (i denotes an integer that satisfies 0≦i≦N−1))to achieve a period (cycle) of N (value reached given that FIGS. 3, 4,6, 12, 25, 29, 51, and 53 perform a change of phase on only one basebandsignal). The present description discusses performing a change of phaseon one precoded baseband signal (i.e., in FIGS. 3, 4, 6, 12, 25, 29, and51) namely on precoded baseband signal z2′. Here, PHASE[k] is calculatedas follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 51} \right\rbrack & \; \\{{{PHASE}\lbrack k\rbrack} = {\frac{2k\; \pi}{N}{radians}}} & \left( {{formula}\mspace{14mu} 51} \right)\end{matrix}$

where k=0, 1, 2 . . . N−2, N−1 (k denotes an integer that satisfies0≦k≦N−1). When N=5, 7, 9, 11, or 15, the reception device is able toobtain good data reception quality.

Although the present description discusses the details of phase changingschemes involving two modulated signals transmitted by a plurality ofantennas, no limitation is intended in this regard. Precoding and achange of phase may be performed on three or more baseband signals onwhich mapping has been performed according to a modulation scheme,followed by predetermined processing on the post-phase change basebandsignals and transmission using a plurality of antennas, to realize thesame results.

Programs for executing the above transmission scheme may, for example,be stored in advance in ROM (Read-Only Memory) and be read out foroperation by a CPU.

Furthermore, the programs for executing the above transmission schememay be stored on a computer-readable recording medium, the programsstored in the recording medium may be loaded in the RAM (Random AccessMemory) of the computer, and the computer may be operated in accordancewith the programs.

The components of the above-described Embodiments may be typicallyassembled as an LSI (Large Scale Integration), a type of integratedcircuit. Individual components may respectively be made into discretechips, or a subset or entirety of the components may be made into asingle chip. Although an LSI is mentioned above, the terms IC(Integrated Circuit), system LSI, super LSI, or ultra LSI may alsoapply, depending on the degree of integration. Furthermore, the methodof integrated circuit assembly is not limited to LSI. A dedicatedcircuit or a general-purpose processor may be used. After LSI assembly,a FPGA (Field Programmable Gate Array) or reconfigurable processor maybe used.

Furthermore, should progress in the field of semiconductors or emergingtechnologies lead to replacement of LSI with other integrated circuitmethods, then such technology may of course be used to integrate thefunctional blocks. Applications to biotechnology are also plausible.

INDUSTRIAL APPLICABILITY

The present invention is widely applicable to wireless systems thattransmit a plurality of different modulated signals from a plurality ofantennas, such as an OFDM-MIMO system. Furthermore, in a wiredcommunication system with a plurality of transmission locations (such asa PLC (Power Line Communication) system, optical communication system,or DSL (Digital Subscriber Line) system), the present invention may beadapted to a MIMO system, where a plurality of transmission locationsare used to transmit a plurality of modulated signals as described bythe present invention. Modulated signals may also be transmitted from aplurality of transmission locations.

REFERENCE SIGNS LIST

-   -   302A, 302B Encoders    -   304A, 304B Interleavers    -   306A, 306B Mappers    -   314 Signal processing scheme information generator    -   308A, 308B Weighting units    -   310A, 310B Wireless units    -   312A, 312B Antennas    -   317A, 317B Phase changers    -   402 Encoder    -   404 Distributor    -   504#1, 504#2 Transmit antennas    -   505#1, 505#2 Receive antennas    -   600 Weighting unit    -   701_X, 701_Y Antennas    -   703_X, 703_Y Wireless units    -   705_1 Channel fluctuation estimator    -   705_2 Channel fluctuation estimator    -   707_1 Channel fluctuation estimator    -   707_2 Channel fluctuation estimator    -   709 Control information decoder    -   711 Signal processor    -   803 Inner MIMO detector    -   805A, 805B Log-likelihood calculators    -   807A, 807B Deinterleavers    -   809A, 809B Log-likelihood ratio calculators    -   811A, 811B Soft-in/soft-out decoders    -   813A, 813B Interleavers    -   815 Memory    -   819 Coefficient generator    -   901 Soft-in/soft-out decoder    -   903 Distributor    -   1201A, 1201B OFDM-related processors    -   1302A, 1302A Serial-to-parallel converters    -   1304A, 1304B Reorderers    -   1306A, 1306B IFFT units    -   1308A, 1308B Wireless units

1. A broadcast signal generation method by a broadcast apparatus,comprising the steps of: applying a phase change to a first basebandsignal generated from a first set of bits and a second baseband signalgenerated from a second set of bits; and applying a precoding to thefirst baseband signal and the second baseband signal according to adetermined matrix F, the precoded first baseband signal and the precodedsecond baseband signal being outputted to a plurality of transmissionantennas to be transmitted on a same frequency band and at a same time,wherein the phase change is continually applied to the first basebandsignal and the second baseband signal using a phase change valuesequentially selected from among N phase change values, N being aninteger greater than two and greater than the number of basebandsignals, and each of the N phase change values being selected at leastonce within a determined period, and a difference between two adjacentphase change values of the N phase change values is 2π/N.
 2. Thebroadcast signal generation method according to claim 1, wherein theprecoding satisfies the relation:(z1,z2)^(T) =F(s1,s2)^(T) wherein z1 and z2 are signals after theprecoding, s1 and s2 are signals before the precoding, (z1, z2) is a rowvector composed of the signals z1 and z2, (z1, z2)^(T) is a transposevector of the row vector (z1, z2), (s1, s2) is a row vector composed ofthe signals s1 and s2, (s1, s2)^(T) is a transpose vector of the rowvector (s1, s2), and F(s1, s2)^(T) is a matrix product of the determinedmatrix F and the transpose vector (s1, s2)^(T).
 3. A broadcast apparatuscomprising: phase changing circuitry applying a phase change to a firstbaseband signal generated from a first set of bits and a second basebandsignal generated from a second set of bits; and precoding circuitryapplying a precoding to the first baseband signal and the secondbaseband signal according to a determined matrix F, the precoded firstbaseband signal and the precoded second baseband signal being outputtedto a plurality of transmission antennas to be transmitted on a samefrequency band and at a same time, wherein the phase change iscontinually applied to the first baseband signal and the second basebandsignal using a phase change value sequentially selected from among Nphase change values, N being an integer greater than two and greaterthan the number of baseband signals, each of the N phase change valuesbeing selected at least once within a determined period, and adifference between two adjacent phase change values of the N phasechange values is 2π/N.
 4. The broadcast apparatus according to claim 3,wherein the precoding satisfies the relation:(z1,z2)^(T) =F(s1,s2)^(T) wherein z1 and z2 are signals after theprecoding, s1 and s2 are signals before the precoding, (z1, z2) is a rowvector composed of the signals z1 and z2, (z1, z2)^(T) is a transposevector of the row vector (z1, z2), (s1, s2) is a row vector composed ofthe signals s1 and s2, (s1, s2)^(T) is a transpose vector of the rowvector (s1, s2), and F(s1, s2)^(T) is a matrix product of the determinedmatrix F and the transpose vector (s1, s2)^(T).
 5. A broadcast signalreception method comprising the steps of: obtaining a reception signal,the reception signal being obtained by receiving a plurality oftransmission signals with a reception antenna, the plurality oftransmission signals being transmitted on a same frequency band and at asame time from a broadcast apparatus with a plurality of transmissionantennas; and demodulating the reception signal in accordance with aphase change value sequentially selected from among N phase changevalues, the modulated reception signal being output to a display,wherein the transmission signals are generated by applying a phasechange to a first baseband signal generated from a first set of bits anda second baseband signal generated from a second set of bits, andapplying a precoding to the first baseband signal and the secondbaseband signal according to a determined matrix F, the phase change iscontinually applied to the first baseband signal and the second basebandsignal using the sequentially selected phase change value, N being aninteger greater than two and greater than the number of basebandsignals, each of the N phase change values being selected at least oncewithin a determined period and a difference between two adjacent phasechange values of the N phase change values is 2π/N.
 6. The broadcastsignal reception method according to claim 5, wherein the precodingsatisfies the relation:(z1,z2)^(T) =F(s1,s2)^(T) wherein z1 and z2 are signals after theprecoding, s1 and s2 are signals before the precoding, (z1, z2) is a rowvector composed of the signals z1 and z2, (z1, z2)^(T) is a transposevector of the row vector (z1, z2), (s1, s2) is a row vector composed ofthe signals s1 and s2, (s1, s2)^(T) is a transpose vector of the rowvector (s1, s2), and F(s1, s2)^(T) is a matrix product of the determinedmatrix F and the transpose vector (s1, s2)^(T).
 7. A broadcast signalreception apparatus comprising: obtaining circuitry obtaining areception signal, the reception signal being obtained by receiving aplurality of transmission signals with a reception antenna, theplurality of transmission signals being transmitted on a same frequencyband and at a same time from a broadcast apparatus with a plurality oftransmission antennas; and demodulating circuitry demodulating thereception signal in accordance with a phase change value sequentiallyselected from among N phase change values, the modulated receptionsignal being outputted to a display, wherein the transmission signalsare generated by applying a phase change to a first baseband signalgenerated from a first set of bits and a second baseband signalgenerated from a second set of bits, and applying a precoding to thefirst baseband signal and the second baseband signal according to adetermined matrix F, the phase change is continually applied to thefirst baseband signal and the second baseband signal using thesequentially selected phase change value, N being an integer greaterthan two and greater than the number of baseband signals, each of the Nphase change values being selected at least once within a determinedperiod, and a difference between two adjacent phase change values of theN phase change values is 2π/N.
 8. The broadcast signal receptionapparatus according to claim 7, wherein the precoding satisfies therelation:(z1,z2)^(T) =F(s1,s2)^(T) wherein z1 and z2 are signals after theprecoding, s1 and s2 are signals before the precoding, (z1, z2) is a rowvector composed of the signals z1 and z2, (z1, z2)^(T) is a transposevector of the row vector (z1, z2), (s1, s2) is a row vector composed ofthe signals s1 and s2, (s1, s2)^(T) is a transpose vector of the rowvector (s1, s2), and F(s1, s2)^(T) is a matrix product of the determinedmatrix F and the transpose vector (s1, s2)^(T).
 9. The broadcast signalgeneration method according to claim 1, wherein the precoding is appliedafter applying the phase change.
 10. The broadcast apparatus accordingto claim 3, wherein the precoding is applied after applying the phasechange.
 11. The broadcast signal reception method according to claim 5,wherein the precoding is applied after applying the phase change. 12.The broadcast signal reception apparatus according to claim 7, whereinthe precoding is applied after applying the phase change.
 13. Thebroadcast signal generation method according to claim 1, wherein thephase change is applied after applying the precoding.
 14. The broadcastapparatus according to claim 3, wherein the phase change is appliedafter applying the precoding.
 15. The broadcast signal reception methodaccording to claim 5, wherein the phase change is applied after applyingthe precoding.
 16. The broadcast signal reception apparatus according toclaim 7, wherein the phase change is applied after applying theprecoding.